Method and apparatus for use in improving linearity of MOSFETs using an accumulated charge sink

ABSTRACT

A method and apparatus for use in improving the linearity characteristics of MOSFET devices using an accumulated charge sink (ACS) are disclosed. The method and apparatus are adapted to remove, reduce, or otherwise control accumulated charge in SOI MOSFETs, thereby yielding improvements in FET performance characteristics. In one exemplary embodiment, a circuit having at least one SOI MOSFET is configured to operate in an accumulated charge regime. An accumulated charge sink, operatively coupled to the body of the SOI MOSFET, eliminates, removes or otherwise controls accumulated charge when the FET is operated in the accumulated charge regime, thereby reducing the nonlinearity of the parasitic off-state source-to-drain capacitance of the SOI MOSFET. In RF switch circuits implemented with the improved SOI MOSFET devices, harmonic and intermodulation distortion is reduced by removing or otherwise controlling the accumulated charge when the SOI MOSFET operates in an accumulated charge regime.

This application is a continuation application of co-pending andcommonly assigned U.S. application Ser. No. 16/054,959, “Method andApparatus for use in Improving Linearity of MOSFETs using an AccumulatedCharge Sink”, filed Aug. 3, 2018; which is a continuation application ofcommonly assigned U.S. application Ser. No. 15/707,970, “Method andApparatus for use in Improving Linearity of MOSFETs using an AccumulatedCharge Sink”, filed Sep. 18, 2017, issued on Dec. 11, 2018 as U.S. Pat.No. 10,153,763; application Ser. No. 15/707,970 is a continuationapplication of commonly assigned U.S. application Ser. No. 14/845,154,“Method and Apparatus for use in Improving Linearity of MOSFETs using anAccumulated Charge Sink”, filed Sep. 3, 2015, issued on Oct. 3, 2017 asU.S. Pat. No. 9,780,775; application Ser. No. 14/845,154 is acontinuation application of and commonly assigned U.S. application Ser.No. 13/850,251, “Method and Apparatus for use in Improving Linearity ofMOSFETs using an Accumulated Charge Sink”, filed Mar. 25, 2013, issuedon Sep. 8, 2015 as U.S. Pat. No. 9,130,564, which application Ser. No.13/850,251 is a continuation application of commonly assigned U.S.application Ser. No. 13/412,529, “Method and Apparatus for use inImproving Linearity of MOSFETs Using an Accumulated Charge Sink”, filedMar. 5, 2012, issued on Mar. 26, 2013 as U.S. Pat. No. 8,405,147, whichapplication Ser. No. 13/412,529 is a Continuation of commonly assignedU.S. application Ser. No. 13/053,211, “Method and Apparatus for use inImproving Linearity of MOSFETs Using an Accumulated Charge Sink”, filedMar. 22, 2011, issued Mar. 6, 2012 as U.S. Pat. No. 8,129,787, whichapplication Ser. No. 13/053,211 is a divisional application of commonlyassigned U.S. application Ser. No. 11/484,370, “Method and Apparatus foruse in Improving Linearity of MOSFETs Using an Accumulated Charge Sink”,filed Jul. 10, 2006, issued Mar. 22, 2011 as U.S. Pat. No. 7,910,993;and application Ser. No. 11/484,370 (U.S. Pat. No. 7,910,993) claims thebenefit of priority under 35 U.S.C. § 119(e) to commonly-assigned U.S.Provisional Application No. 60/698,523, filed Jul. 11, 2005, entitled“Method and Apparatus for use in Improving Linearity of MOSFETs using anAccumulated Charge Sink”; application Ser. No. 16/054,959 is also acontinuation of commonly assigned pending U.S. application Ser. No.15/419,898 filed Jan. 30, 2017, which is a continuation application ofcommonly assigned U.S. application Ser. No. 13/948,094 filed Jul. 22,2013 (U.S. Pat. No. 9,608,619 issued on Mar. 28, 2017), which is acontinuation application of commonly assigned U.S. application Ser. No.13/028,144 filed Feb. 15, 2011 (U.S. Pat. No. 8,954,902 issued on Feb.10, 2015), which is a divisional of commonly assigned U.S. applicationSer. No. 11/520,912 filed Sep. 14, 2006 (U.S. Pat. No. 7,890,891 issuedon Feb. 15, 2011), which is a continuation-in-part of U.S. applicationSer. No. 11/484,370, filed Jul. 10, 2006 (U.S. Pat. No. 7,910,993 issuedon Mar. 22, 2011), and which Ser. No. 11/520,912 application claimspriority to U.S. provisional applications 60/718,260 filed Sep. 15,2005; and this Continuation application is also related to U.S.application Ser. No. 11/881,816 filed Jul. 26, 2007 which is a CIP ofSer. No. 11/520,912 and a CIP of Ser. No. 11/484,370; and the contentsof all of the above cited provisional applications, pendingapplications, and issued patents, including their associated appendices,are hereby incorporated by reference herein in their entirety.

BACKGROUND

1. Field

The present invention relates to metal-oxide-semiconductor (MOS) fieldeffect transistors (FETs), and particularly to MOSFETs fabricated onSemiconductor-On-Insulator (“SOI”) and Semiconductor-On-Sapphire (“SOS”)substrates. In one embodiment, an SOI (or SOS) MOSFET is adapted tocontrol accumulated charge and thereby improve linearity of circuitelements.

2. Description of Related Art

Although the disclosed method and apparatus for use in improving thelinearity of MOSFETs are described herein as applicable for use in SOIMOSFETs, it will be appreciated by those skilled in the electronicdevice design arts that the present teachings are equally applicable foruse in SOS MOSFETs. In general, the present teachings can be used in theimplementation of MOSFETs using any convenientsemiconductor-on-insulator technology, including silicon-on-insulatortechnology. For example, the inventive MOSFETs described herein can beimplemented using compound semiconductors on insulating substrates. Suchcompound semiconductors include, but are not limited to, the following:Silicon Germanium (SiGe), Gallium Arsenide (GaAs), Indium Phosphide(InP), Gallium Nitride (GaN), Silicon Carbide (SiC), and II-VI compoundsemiconductors, including Zinc Selenide (ZnSe) and Zinc Sulfide (ZnS).The present teachings also may be used in implementing MOSFETsfabricated from thin-film polymers. Organic thin-film transistors(OTFTs) utilize a polymer, conjugated polymers, oligomers, or othermolecules to form the insulting gate dielectric layer. The presentinventive methods and apparatus may be used in implementing such OTFTs.

It will be appreciated by those skilled in the electronic design artsthat the present disclosed method and apparatus apply to virtually anyinsulating gate technology, and to integrated circuits having a floatingbody. As those skilled in the art will appreciate, technologies areconstantly being developed for achieving “floating body”implementations. For example, the inventors are aware of circuitsimplemented in bulk silicon wherein circuit implementations are used to“float” the body of the device. In addition, the disclosed method andapparatus can also be implemented using silicon-on-bonded waferimplementations. One such silicon-on-bonded wafer technique uses “directsilicon bonded” (DSB) substrates. Direct silicon bond (DSB) substratesare fabricated by bonding and electrically attaching a film ofsingle-crystal silicon of differing crystal orientation onto a basesubstrate. The present disclosure therefore contemplates embodiments ofthe disclosed method and apparatus implemented in any of the developingfloating body implementations. Therefore, references to and exemplarydescriptions of SOI MOSFETs herein are not to be construed as limitingthe applicability of the present teachings to SOI MOSFETs only. Rather,as described below in more detail, the disclosed method and apparatusfind utility in MOSFETs implemented in a plurality of devicetechnologies, including SOS and silicon-on-bonded wafer technologies.

As is well known, a MOSFET employs a gate-modulated conductive channelof n-type or p-type conductivity, and is accordingly referred to as an“NMOSFET” or “PMOSFET”, respectively. FIG. 1 shows a cross-sectionalview of an exemplary prior art SOI NMOSFET 100. As shown in FIG. 1, theprior art SOI NMOSFET 100 includes an insulating substrate 118 that maycomprise a buried oxide layer, sapphire, or other insulating material. Asource 112 and drain 116 of the NMOSFET 100 comprise N+ regions (i.e.,regions that are heavily doped with an “n-type” dopant material)produced by ion implantation into a silicon layer positioned above theinsulating substrate 118. (The source and drain of PMOSFETs comprise P+regions (i.e., regions heavily doped with “p-type” dopant material)).The body 114 comprises a P− region (i.e., a region that is lightly dopedwith a “p-type” dopant), produced by ion implantation, or by dopantsalready present in the silicon layer when it is formed on the insulatingsubstrate 118. As shown in FIG. 1, the NMOSFET 100 also includes a gateoxide 110 positioned over the body 114. The gate oxide 110 typicallycomprises a thin layer of an insulating dielectric material such asSiO₂. The gate oxide 110 electrically insulates the body 114 from a gate108 positioned over the gate oxide 110. The gate 108 comprises a layerof metal or, more typically, polysilicon.

A source terminal 102 is operatively coupled to the source 112 so that asource bias voltage “Vs” may be applied to the source 112. A drainterminal 106 is operatively coupled to the drain 116 so that a drainbias voltage “Vd” may be applied to the drain 116. A gate terminal 104is operatively coupled to the gate 108 so that a gate bias voltage “Vg”may be applied to the gate 108.

As is well known, when a voltage is applied between the gate and sourceterminals of a MOSFET, a generated electric field penetrates through thegate oxide to the transistor body. For an enhancement mode device, apositive gate bias creates a channel in the channel region of the MOSFETbody through which current passes between the source and drain. For adepletion mode device, a channel is present for a zero gate bias.Varying the voltage applied to the gate modulates the conductivity ofthe channel and thereby controls the current flow between the source anddrain.

For an enhancement mode MOSFET, for example, the gate bias creates aso-called “inversion channel” in a channel region of the body 114 underthe gate oxide 110. The inversion channel comprises carriers having thesame polarity (e.g., “P” polarity (i.e., hole carriers), or “N” polarity(i.e., electron carriers) carriers) as the polarity of the source anddrain carriers, and it thereby provides a conduit (i.e., channel)through which current passes between the source and the drain. Forexample, as shown in the SOI NMOSFET 100 of FIG. 1, when a sufficientlypositive voltage is applied between the gate 108 and the source 112(i.e. a positive gate bias exceeding a threshold voltage V_(th)), aninversion channel is formed in the channel region of the body 114. Asnoted above, the polarity of carriers in the inversion channel isidentical to the polarity of carriers in the source and drain. In thisexample, because the source and drain comprise “n-type” dopant materialand therefore have N polarity carriers, the carriers in the channelcomprise N polarity carriers. Similarly, because the source and draincomprise “p-type” dopant material in PMOSFETs, the carriers in thechannel of turned on (i.e., conducting) PMOSFETs comprise P polaritycarriers.

Depletion mode MOSFETs operate similarly to enhancement mode MOSFETs,however, depletion mode MOSFETs are doped so that a conducting channelexists even without a voltage being applied to the gate. When a voltageof appropriate polarity is applied to the gate the channel is depleted.This, in turn, reduces the current flow through the depletion modedevice. In essence, the depletion mode device is analogous to a“normally closed” switch, while the enhancement mode device is analogousto a “normally open” switch. Both enhancement and depletion mode MOSFETshave a gate voltage threshold, V_(th), at which the MOSFET changes froman off-state (non-conducting) to an on-state (conducting).

No matter what mode of operation an SOI MOSFET employs (i.e., whetherenhancement or depletion mode), when the MOSFET is operated in anoff-state (i.e., the gate voltage does not exceed V_(th)), and when asufficient nonzero gate bias voltage is applied with respect to thesource and drain, an “accumulated charge” may occur under the gate. The“accumulated charge”, as defined in more detail below and usedthroughout the present application, is similar to the “accumulationcharge” described in the prior art literature in reference to MOScapacitors. However, the prior art references describe “accumulationcharge” as referring only to bias-induced charge existing under a MOScapacitor oxide, wherein the accumulation charge is of the same polarityas the majority carriers of the semiconductor material under thecapacitor oxide. In contrast, and as described below in more detail,“accumulated charge” is used herein to refer to gate-bias inducedcarriers that may accumulate in the body of an off-state MOSFET, even ifthe majority carriers in the body do not have the same polarity as theaccumulated charge. This situation may occur, for example, in anoff-state depletion mode NMOSFET, wherein the accumulated charge maycomprise holes (i.e., having P polarity) even though the body doping isN− rather than P−.

For example, as shown in FIG. 1, when the SOI NMOSFET 100 is biased tooperate in an off-state, and when a sufficient nonzero voltage isapplied to the gate 108, an accumulated charge 120 may accumulate in thebody 114 underneath and proximate the gate oxide 110. The operatingstate of the SOI NMOSFET 100 shown in FIG. 1 is referred to herein as an“accumulated charge regime” of the MOSFET. The accumulated charge regimeis defined in more detail below. The causes and effects of theaccumulated charge in SOI MOSFETs are now described in more detail.

As is well known, electron-hole pair carriers may be generated in MOSFETbodies as a result of several mechanisms (e.g., thermal, optical, andband-to-band tunneling electron-hole pair generation processes). Whenelectron-hole pair carriers are generated within an NMOSFET body, forexample, and when the NMOSFET is biased in an off-state condition,electrons may be separated from their hole counterparts and pulled intoboth the source and drain. Over a period of time, assuming the NMOSFETcontinues to be biased in the off-state, the holes (resulting from theseparated electron-hole pairs) may accumulate under the gate oxide(i.e., forming an “accumulated charge”) underneath and proximate thegate oxide. A similar process (with the behavior of electrons and holesreversed) occurs in similarly biased PMOSFET devices. This phenomenon isnow described with reference to the SOI NMOSFET 100 of FIG. 1.

When the SOI NMOSFET 100 is operated with gate, source and drain biasvoltages that deplete the channel carriers in the body 114 (i.e., theNMOSFET 100 is in the off-state), holes may accumulate underneath andproximate the gate oxide 110. For example, if the source bias voltage Vsand the drain bias voltage Vd are both zero (e.g., connected to a groundcontact, not shown), and the gate bias voltage Vg comprises asufficiently negative voltage with respect to ground and with respect toV_(th), holes present in the body 114 become attracted to the channelregion proximate the gate oxide 110. Over a period of time, unlessremoved or otherwise controlled, the holes accumulate underneath thegate oxide 110 and result in the accumulated charge 120 shown in FIG. 1.The accumulated charge 120 is therefore shown as positive “+” holecarriers in FIG. 1. In the example given, Vg is negative with respect toVs and Vd, so electric field regions 122 and 124 may also be present.

Accumulated Charge Regime Defined

The accumulated charge is opposite in polarity to the polarity ofcarriers in the channel. Because, as described above, the polarity ofcarriers in the channel is identical to the polarity of carriers in thesource and drain, the polarity of the accumulated charge 120 is alsoopposite to the polarity of carriers in the source and drain. Forexample, under the operating conditions described above, holes (having“P” polarity) accumulate in off-state NMOSFETs, and electrons (having“N” polarity) accumulate in off-state PMOSFETs. Therefore, a MOSFETdevice is defined herein as operating within the “accumulated chargeregime” when the MOSFET is biased to operate in an off-state, and whencarriers having opposite polarity to the channel carriers are present inthe channel region. Stated in other terms, a MOSFET is defined asoperating within the accumulated charge regime when the MOSFET is biasedto operate in an off-state, and when carriers are present in the channelregion having a polarity that is opposite the polarity of the source anddrain carriers.

For example, and referring again to FIG. 1, the accumulated charge 120comprises hole carriers having P or “+” polarity. In contrast, thecarriers in the source, drain, and channel (i.e., when the FET is in theon-state) comprise electron carriers having N or “−” polarity. The SOINMOSFET 100 is therefore shown in FIG. 1 as operating in the accumulatedcharge regime. It is biased to operate in an off-state, and anaccumulated charge 120 is present in the channel region. The accumulatedcharge 120 is opposite in polarity (P) to the polarity of the channel,source and drain carriers (N).

In another example, wherein the SOI NMOSFET 100 comprises a depletionmode device, V_(th) is negative by definition. According to thisexample, the body 114 comprises an N− region (as contrasted with the P−region shown in FIG. 1). The source and drain comprise N+ regionssimilar to those shown in the enhancement mode MOSFET 100 of FIG. 1. ForVs and Vd both at zero volts, when a gate bias Vg is applied that issufficiently negative relative to V_(th) (for example, a Vg that is morenegative than approximately −1 V relative to V_(th)), the depletion modeNMOSFET is biased into an off-state. If biased in the off-state for asufficiently long period of time, holes may accumulate under the gateoxide and thereby comprise the accumulated charge 120 shown in FIG. 1.

In other examples, Vs and Vd may comprise nonzero bias voltages. In someembodiments, Vg must be sufficiently negative to both Vs and Vd (inorder for Vg to be sufficiently negative to V_(th), for example) inorder to bias the NMOSFET in the off-state. Those skilled in the MOSFETdevice design arts shall recognize that a wide variety of bias voltagesmay be used to practice the present teachings. As described below inmore detail, the present disclosed method and apparatus contemplates usein any SOI MOSFET device biased to operate in the accumulated chargeregime.

SOI and SOS MOSFETs are often used in applications in which operationwithin the accumulated charge regime adversely affects MOSFETperformance. As described below in more detail, unless the accumulatedcharge is removed or otherwise controlled, it detrimentally affectsperformance of SOI MOSFETs under certain operating conditions. Oneexemplary application, described below in more detail with reference tothe circuits shown in FIGS. 2B and 5A, is the use of SOI MOSFETs in theimplementation of radio frequency (RF) switching circuits. As describedbelow with reference to FIGS. 2B and 5A in more detail, the inventorshave discovered that unless the accumulated charge is removed orotherwise controlled, under some operating conditions, the accumulatedcharge adversely affects the linearity of the SOI MOSFET and therebyincreases harmonic distortion and intermodulation distortion (IMD)caused by the MOSFET when used in the implementation of certaincircuits. In addition, as described below in more detail, the inventorshave discovered that removal or control of the accumulated chargeimproves the drain-to-source breakdown voltage (i.e., the “BVDSS”)characteristics of the SOI MOSFETs.

Therefore, it is desirable to provide techniques for adapting andimproving SOI (and SOS) MOSFETs, and circuits implemented with theimproved SOI MOSFETs, in order to remove or otherwise control theaccumulated charge, and thereby significantly improve SOI MOSFETperformance. It is desirable to provide methods and apparatus for use inimproving the linearity characteristics in SOI MOSFETs. The improvedMOSFETs should have improved linearity, harmonic distortion,intermodulation distortion, and BVDSS characteristics as compared withprior art MOSFETs, and thereby improve the performance of circuitsimplemented with the improved MOSFETs. The present teachings providesuch novel methods and apparatus.

SUMMARY

Apparatuses and methods are provided to control accumulated charge inSOI MOSFETs, thereby improving nonlinear responses and harmonic andintermodulaton distortion effects in the operation of the SOI MOSFETs.

In one embodiment, a circuit having at least one SOI MOSFET isconfigured to operate in an accumulated charge regime. An accumulatedcharge sink (ACS), operatively coupled to the body of the SOI MOSFET,receives accumulated charge generated in the body, thereby reducing thenonlinearity of the net source-drain capacitance of the SOI MOSFET.

In one embodiment, the ACS comprises a high impedance connection to theMOSFET body, with an exemplary impedance greater than 10⁶ ohm.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a cross-sectional view of an exemplary prior art SOI NMOSFET.

FIG. 2A is a simplified schematic of an electrical model showing theoff-state impedance characteristics of the exemplary prior art SOINMOSFET of FIG. 1.

FIG. 2B is a schematic of an exemplary simplified RF switching circuitimplemented using prior art SOI MOSFETs such as the prior art SOINMOSFET of FIG. 1.

FIGS. 3A and 3B are simplified schematic diagrams of a top view of animproved SOI NMOSFET adapted to control accumulated charge in accordancewith the present teachings.

FIG. 3C is a cross-sectional perspective schematic of an improved SOINMOSFET adapted to control accumulated charge showing gate, source,drain and accumulated charge sink (ACS) terminals.

FIG. 3D is a simplified top view schematic of an improved SOI NMOSFETadapted to control accumulated charge having an accumulated charge sink(ACS) electrically coupled to a P+ region.

FIG. 3E is a simplified top view schematic of an improved SOI NMOSFETadapted to control accumulated charge and showing a cross-sectional viewline A-A′ taken along approximately a center of the SOI NMOSFET.

FIG. 3F is a cross-sectional view of the improved SOI NMOSET of FIG. 3Etaken along the A-A′ view line of FIG. 3E.

FIG. 3G is a cross-sectional view of the improved SOI NMOSET of FIGS.3A-3B.

FIG. 3H is a simplified top view schematic of an SOI NMOSFETillustrating a region of increased threshold voltage that can occur inprior art MOSFETs and in some embodiments of the improved SOI MOSFET dueto manufacturing processes.

FIG. 3I is a plot of inversion channel charge as a function of appliedgate voltage when a region of increased threshold voltage is present inan SOI MOSFET.

FIG. 3J is a simplified top view schematic of an improved SOI NMOSFETadapted to control accumulated charge and configured in a “T-gate”configuration.

FIG. 3K is a simplified top view schematic of an improved SOI NMOSFETadapted to control accumulated charge and configured in an “H-gate”configuration.

FIG. 4A is a simplified schematic of an improved SOI NMOSFET adapted tocontrol accumulated charge embodied as a four terminal device.

FIG. 4B is a simplified schematic of an improved SOI NMOSFET adapted tocontrol accumulated charge, embodied as a four terminal device, whereinan accumulated charge sink (ACS) terminal is coupled to a gate terminal.

FIG. 4C is a simplified schematic of an improved SOI NMOSFET adapted tocontrol accumulated charge, embodied as a four terminal device, whereinan accumulated charge sink (ACS) terminal is coupled to a gate terminalvia a diode.

FIG. 4D is a simplified schematic of an improved SOI NMOSFET adapted tocontrol accumulated charge, embodied as a four terminal device, whereinan accumulated charge sink (ACS) terminal is coupled to a controlcircuit.

FIG. 4E is a simplified schematic of an exemplary RF switch circuitimplemented using the four terminal ACC NMOSFET of FIG. 4D, wherein theACS terminal is driven by an external bias source.

FIG. 4F is a simplified schematic of an improved SOI NMOSFET adapted tocontrol accumulated charge, embodied as a four terminal device, whereinan accumulated charge sink (ACS) terminal is coupled to a clampingcircuit.

FIG. 4G is a simplified schematic of an improved SOI NMOSFET adapted tocontrol accumulated charge, embodied as a four terminal device, whereinan accumulated charge sink (ACS) terminal is coupled to a gate terminalvia a diode in parallel with a capacitor.

FIG. 4H shows plots of the off-state capacitance (C_(off)) versusapplied drain-to-source voltages for SOI MOSFETs operated in theaccumulated charge regime, wherein a first plot shows the off-statecapacitance C_(off) of a prior art SOI MOSFET, and wherein a second plotshows the off-state capacitance C_(off) of the improved ACC SOI MOSFETmade in accordance with the present teachings.

FIG. 5A is a schematic of an exemplary prior art single pole, singlethrow (SPST) radio frequency (RF) switch circuit.

FIG. 5B is a schematic of an RF switch circuit adapted for improvedperformance using accumulated charge control, wherein the gate of ashunting SOI NMOSFET is coupled to an accumulated charge sink (ACS)terminal.

FIG. 5C is a schematic of an RF switch circuit adapted for improvedperformance using accumulated charge control, wherein the gate of ashunting SOI NMOSFET is coupled to an accumulated charge sink (ACS)terminal via a diode.

FIG. 5D is a schematic of an RF switch circuit adapted for improvedperformance using accumulated charge control, wherein the accumulatedcharge sink (ACS) terminal is coupled to a control circuit.

FIG. 6 is a schematic of an RF switch circuit including stacked MOSFETs,adapted for improved performance using accumulated charge control,wherein the accumulated charge sink (ACS) terminals of the shuntingstacked MOSFETs are coupled to a control signal.

FIG. 7 shows a flowchart of an exemplary method of improving thelinearity of an SOI MOSFET device using an accumulated charge sink inaccordance with the present disclosure.

FIG. 8 shows a simplified circuit schematic of an exemplary embodimentof an RF switch circuit made in accordance with the present disclosure,wherein the RF switch circuit includes drain-to-source resistors betweenthe drain and source of the ACC MOSFETs.

FIG. 9 shows a simplified schematic of an exemplary single-poledouble-throw (SPDT) RF switch circuit made in accordance with thepresent disclosure, wherein drain-to-source resistors are shown acrossthe switching ACC SOI MOSFETs.

Like reference numbers and designations in the various drawings indicatelike elements.

DETAILED DESCRIPTION

As noted above, those skilled in the electronic device design arts shallappreciate that the teachings herein apply equally to NMOSFETs andPMOSFETs. For simplicity, the embodiments and examples presented hereinfor illustrative purposes include only NMOSFETs, unless otherwise noted.By making well known changes to dopants, charge carriers, polarity ofbias voltages, etc., persons skilled in the arts of electronic deviceswill easily understand how these embodiments and examples may be adaptedfor use with PMOSFETs.

Non-linearity and Harmonic Distortion Effects of Accumulated Charge inan SOI NMOSFET

As described above in the background, no matter what mode of operationthe MOSFET employs (i.e., enhancement mode or depletion mode), undersome circumstances, when a MOSFET is operated in an off-state with anonzero gate bias voltage applied with respect to the source and drain,an accumulated charge may occur under the gate. According to the presentteachings, as described above when the MOSFET is in an off-state, andwhen carriers are present in the channel region having a polarity thatis opposite the polarity of the source and drain carriers, the MOSFET isdefined herein as operating in the accumulated charge regime.

According to the present teachings, the inventors have observed that,when used in certain circuit implementations, MOSFETs operating in theaccumulated charge regime exhibit undesirable non-linear characteristicsthat adversely impact circuit performance. For example, as describedbelow in more detail with reference to FIG. 2A, the accumulated charge120 (FIG. 1) adversely affects the linearity of off-state SOI MOSFETs,and more specifically, it adversely affects the linearity ofcontributing capacitances to the drain-to-source capacitance (Cds). Foran SOI MOSFET operating in an off-state, Cds is referred to as C_(off).The contributing capacitances to C_(off) are described below inreference to FIG. 2A for bias conditions wherein the gate bias Vg isprovided by a circuit having an impedance that is large compared to theimpedances of the contributing capacitances. As described below withreference to FIGS. 2B and 5A, this, in turn, adversely affects harmonicdistortion, intermodulation distortion, and other performancecharacteristics of circuits implemented with the SOI MOSFETs. Thesenovel observations, not taught or suggested by the prior art, may beunderstood with reference to the electrical model shown in FIG. 2A.

FIG. 2A is a simplified schematic of an electrical model 200 showing theoff-state impedance (or conversely, conductance) characteristics of theexemplary prior art SOI NMOSFET 100 of FIG. 1. More specifically, themodel 200 shows the impedance characteristics from the source 112 to thedrain 116 when the NMOSFET 100 is operated in the off-state. Because thedrain-to-source off-state impedance characteristic of the NMOSFET 100 isprimarily capacitive in nature, it is referred to herein as thedrain-to-source off-state capacitance (C_(off)). For the exemplarydescription herein, the gate 108 is understood to be biased at a voltageVg by a circuit (not shown) that has an impedance that is large comparedto the impedances of the contributing capacitances described inreference to FIG. 2A. Persons skilled in the electronic arts willunderstand how this exemplary description may be modified for the casewherein the impedance of the circuit providing the Vg bias is not largecompared to the impedances of the contributing capacitances.

As shown in FIG. 2A, the junction between the source 112 and the body114 (i.e., a source-body junction 218) of the off-state NMOSFET 100 canbe represented by a junction diode 208 and a junction capacitor 214,configured as shown. Similarly, the junction between the drain 116 andthe body 114 (i.e., the drain-body junction 220) of the off-stateNMOSFET 100 can be represented by a junction diode 210 and a junctioncapacitor 216, configured as shown. The body 114 is represented simplyas an impedance 212 that is present between the source-body junction 218and the drain-body junction 220.

A capacitor 206 represents the capacitance between the gate 108 and thebody 114. A capacitor 202 represents the capacitance between the source112 and the gate 108, and another capacitor 204 represents thecapacitance between the drain 116 and the gate 108. A substratecapacitance due to the electrical coupling between the source 112 andthe drain 116 (through the insulating substrate 118 shown in FIG. 1) istaken to be negligibly small in the exemplary description set forthbelow, and therefore is not shown in the electrical model 200 of FIG.2A.

As described above, when the NMOSFET 100 is in the off-state, and whenthe accumulated charge 120 (FIG. 1) is not present in the body 114(i.e., the NMOSFET 100 is not operating within the accumulated chargeregime), the body 114 is depleted of charge carriers. In this case thebody impedance 212 is analogous to the impedance of an insulator, andthe electrical conductance through the body 114 is very small (i.e., theNMOSFET 100 is in the off-state). Consequently, the principalcontributions to the drain-to-source off-state capacitance C_(off) areprovided by the capacitors 202 and 204. The capacitors 202 and 204 areonly slightly voltage dependent, and therefore do not significantlycontribute to a nonlinear response that adversely affects harmonicgeneration and intermodulation distortion characteristics.

However, when the NMOSFET 100 operates within the accumulated chargeregime, and the accumulated charge 120 is therefore present in the body114, mobile holes comprising the accumulated charge produce p-typeconductivity between the source-body junction 218 and the drain-bodyjunction 220. In effect, the accumulated charge 120 produces animpedance between the source-body junction 218 and the drain-bodyjunction 220 that is significantly less than the impedance between thejunctions in the absence of the accumulated charge. If a Vds voltage isapplied between the drain 116 and the source 112, the mobile holesredistribute according to the electrical potentials that result withinthe body 114. DC and low-frequency current flow through the SOI NMOSFET100 is prevented by the diode properties of the source-body junction 218and the drain-body junction 220, as represented by the junction diodes208 and 210, respectively. That is, because the junction diodes 208 and210 are anti-series (i.e., “back-to-back”) in this case, no DC orlow-frequency currents flow through the SOI NMOSFET 100. However,high-frequency currents may flow through the SOI NMOSFET 100 via thecapacitances of the source-body junction 218 and the drain-body junction220, as represented by the junction capacitors 214 and 216,respectively.

The junction capacitors 214 and 216 are voltage dependent because theyare associated with junctions between n-type and p-type regions. Thisvoltage dependence results from the voltage dependence of the width ofthe depletion region of the junction between the n-type and p-typeregions. As a bias voltage is applied to the NMOSFET, the width of thedepletion region of the junction between the n-type and p-type regionsis varied. Because the capacitance of the junction depends on the widthof the junction depletion region, the capacitance also varies as afunction of the bias applied across the junction (i.e., the capacitanceis also voltage dependent).

Further, the capacitors 202 and 204 may also have a voltage dependencecaused by the presence of the accumulated charge 120. Although thecomplex reasons for this voltage dependence are not described in detailherein, persons skilled in the arts of electronic devices shallunderstand that electric field regions (e.g., electric field regions 122and 124 described above with reference to FIG. 1) may be affected by theresponse of the accumulated charge and its response to an applied Vds,thereby causing a voltage dependence of capacitors 202 and 204. Anadditional nonlinear effect may occur due to a direct capacitance (notshown) between the source 112 and the drain 116. Although this directcapacitance would usually be expected to be negligible for most SOIMOSFETs, it may contribute for SOI MOSFETs having very short spacingbetween the source and drain. The contribution of this directcapacitance to C_(off) is also voltage-dependent in the presence of anaccumulated charge, for reasons that are analogous to the voltagedependencies of the capacitors 202 and 204 as described above.

The voltage dependencies of the junction capacitors 214 and 216, thegate-to-source and gate-to-drain capacitors 202, 204, respectively, andthe direct capacitance (not shown), cause nonlinear behavior inoff-state capacitance C_(off) of the MOSFET when AC voltages are appliedto the NMOSFET 100, thereby producing undesirable generation of harmonicdistortions and intermodulation distortion (IMD). The relativecontributions of these effects are complex, and depend on fabricationprocesses, biases, signal amplitudes, and other variables. However,those skilled in the electronic device design arts shall understand fromthe teachings herein that reducing, removing, or otherwise controllingthe accumulated charge provides an overall improvement in the nonlinearbehavior of C_(off). In addition, because the body impedance 212 issignificantly decreased in the presence of the accumulated charge 120,the magnitude of C_(off) may be increased when the FET operates in theaccumulated charge regime. Reducing, removing, or otherwise controllingthe accumulated charge also mitigates this effect.

In addition, the accumulated charge does not accumulate in the body inan instant as soon as the FET transitions from an on-state (conductingstate) to an off-state (non-conducting state). Rather, when the FETtransitions from the on-state to the off-state, it begins to accumulatecharge in the body of the MOSFET, and the amount of accumulated chargeincreases over time. The accumulation of the accumulated chargetherefore has an associated time constant (i.e., it does not instantlyreach a steady-state level of accumulated charge). The accumulatedcharge accumulates slowly in the FET body. The depleted FET has aC_(off) associated with it which is increased with an increasing amountof accumulated charge. In terms of FET performance, as the C_(off)increases with an increasing amount of accumulated charge in the FETbody, drift occurs in the FET insertion loss (i.e., the FET becomes more“lossy”), isolation (the FET becomes less isolating) and insertion phase(delay in the FET is increased). Reducing, removing, or otherwisecontrolling the accumulated charge also mitigates these undesirabledrift effects.

The inventors have observed that the nonlinear behavior of the MOSFEToff-state capacitance C_(off) adversely affects the performance ofcertain circuits implemented with the prior art SOI MOSFETs. Forexample, when an RF switch is implemented using the prior art SOIMOSFETs, such as the prior art SOI NMOSFET 100 of FIG. 1, theabove-described non-linear off-state characteristics of the prior artMOSFETs adversely affect the linearity of the switch. As described belowin more detail, RF switch linearity is an important design parameter inmany applications. Improved switch linearity leads to improvedsuppression of harmonic and intermodulation (IM) distortion of signalsprocessed by the switch. These improved switch characteristics can becritically important in some applications such as use in cellularcommunication devices.

For example, the well known GSM cellular communication system standardimposes stringent linearity, harmonic and intermodulation suppression,and power consumption requirements on front-end components used toimplement GSM cell phones. One exemplary GSM standard requires that allharmonics of a fundamental signal be suppressed to below −30 dBm atfrequencies up to 12.75 GHz. If harmonics are not suppressed below theselevels, reliable cell phone operation can be significantly adverselyimpacted (e.g., increased dropped calls or other communication problemsmay result due to harmonic and intermodulation distortion of thetransmit and receive signals). Because the RF switching function isgenerally implemented in the cell phone front-end components,improvements in the RF switch linearity, harmonic and intermodulationsuppression, and power consumption performance characteristics is highlydesirable. A description of how the non-linear behavior of the off-statecapacitance C_(off) of the prior art MOSFETs adversely affects these RFswitch characteristics is now described with reference to FIG. 2B.

Harmonic Distortion Effects on RF Switch Circuits Implemented UsingPrior Art SOI MOSFETs

FIG. 2B illustrates an exemplary simplified RF switch circuit 250implemented using prior art MOSFETs such as the prior art SOI NMOSFET100 described above with reference to FIG. 1. A detailed description ofthe operation and implementation of RF switch circuits is provided incommonly assigned U.S. Pat. No. 6,804,502 which is hereby incorporatedherein by reference in its entirety for its teachings on RF switchcircuits. As shown in FIG. 2B, the prior art RF switch 250 includes asingle “pass” or “switching” MOSFET 254 operatively coupled to fiveshunting MOSFETs 260 a-260 e.

The MOSFET 254 acts as a pass or switching transistor and is configured,when enabled, to selectively couple an RF input signal (applied to itsdrain, for example) to an RF antenna 258 via a transmission path 256.The shunting MOSFETs, 260 a-260 e, when enabled, act to alternativelyshunt the RF input signal to ground. As is well known, the switchingMOSFET 254 is selectively controlled by a first switch control signal(not shown) coupled to its gate, and the shunting MOSFETs, 260 a-260 eare similarly controlled by a second switch control signal (not shown)coupled to their gates. The switching MOSFET 254 is thereby enabled whenthe shunting MOSFETs 260 a-260 e are disabled, and vice versa. As shownin the exemplary embodiment of the RF switch 250 of FIG. 2B, theswitching MOSFET 254 is enabled by applying a gate bias voltage of +2.5V(via the first switch control signal). The shunting MOSFETs 260 a-260 eare disabled by applying a gate bias voltage of −2.5V (via the secondswitch control signal).

When the switch 250 is configured in this state, the RF signal 252propagates through the switching MOSFET 254, through the transmissionpath 256, and to the antenna 258. As described above with reference toFIG. 2A, when the shunting MOSFETS 260 a-260 e comprise prior art SOI(or SOS) MOSFETs, such as the SOI NMOSFET 100 (FIG. 1), an accumulatedcharge can occur in the SOI MOSFET bodies (i.e., when the SOI MOSFETsoperate in the accumulated charge regime as described above). Theaccumulated charge can produce nonlinear behavior in the off-statecapacitance C_(off) of the SOI MOSFETs when AC voltages are applied tothe MOSFETs.

More specifically, when the accumulated charge is present in the channelregions of the off-state SOI MOSFETs 260 a-260 e it responds tovariations in the RF signals applied to their respective drains. As thetime varying RF signal propagates along the transmission path 256, theRF signal applies time varying source-to-drain bias voltages to the SOIMOSFETs 260 a-260 e. The time varying source-to-drain bias voltagescreates movement of the accumulated charge within the channel regions ofthe SOI MOSFETs 260-260 e. The movement of the accumulated charge withinthe channel regions of the SOI MOSFETs causes variations in thedrain-to-source off-state capacitance of the SOI MOSFETs 260 a-260 e.More specifically, the movement of the accumulated charge within thechannel regions causes a voltage dependence of the drain-to-sourceoff-state capacitance as described above with reference to FIG. 2A. Thevoltage dependent variations in the off-state capacitance of the SOIMOSFETs 260 a-260 e is the dominant cause of harmonic distortion and IMDof the RF signal as it propagates through the RF switch 250.

As noted above, harmonic distortion and IMD of the RF signal is a majordisadvantage of the prior art RF switch circuits implemented using theprior art SOI MOSFET devices. For many applications, harmonics and IMDof the RF signal must be suppressed to levels that heretofore have beendifficult or impossible to achieve using prior art SOI MOSFET devices.In GSM devices, for example, at a maximum operating power of +35 dBm,prior art switches typically have only a 6 dB margin to the GSM thirdorder harmonics suppression requirement of less than −30 dBm. Very loweven order harmonic distortion is also desirable in GSM systems as thesecond order harmonic of the GSM transmit band also resides in the DCSreceive band. Suppression of odd order (e.g., third order) harmonics ofthe RF signal, however, is desirable and improvements in that regard areneeded.

In addition, as is well known, presence of an accumulated charge in thebodies of floating body (e.g., SOI) MOSFETs can also adversely affectthe drain-to-source breakdown voltage (BVDSS) performancecharacteristics of the floating body MOSFETs. As is well known,floating-body FETs demonstrate drain-to-source breakdown voltageproblems, also known as BVDSS, wherein the drain-to-source“punch-through” voltage is reduced by a parasitic bipolar action. Theparasitic bipolar action is caused when holes are generated in thechannel and the holes have nowhere to dissipate (i.e., because the bodyis floating, the holes have no means for escaping the body). As aconsequence, the potential of the MOSFET body is increased, whicheffectively reduces the threshold voltage. In turn, this conditioncauses the MOSFET device to experience increased leakage, therebygenerating more holes in the body, and thereby exacerbating the BVDSSproblem (as a result of this positive feedback condition).

The present disclosed method and apparatus for improving linearity ofSOI (and SOS) MOSFET devices overcomes the above-described disadvantagesof the prior art. Once the accumulated charge is recognized as a majorsource of harmonic distortion, IMD and compression/saturation inoff-state SOI MOSFET devices, and in circuits (such as RF circuits)implemented with these devices, it becomes clear that reduction,removal, and/or control of the accumulated charge improves the harmonicsuppression characteristics of these devices. In addition, reduction,removal, and/or control of the accumulated charge also improve the BVDSSperformance characteristics by preventing the parasitic bipolar actionfrom occurring. Improvements in BVDSS lead to consequent improvements indevice linearity. Several exemplary structures and techniques forcontrolling the accumulated charge in SOI MOSFETs are described indetail in the next section.

Method and Apparatus for Improving the Linearity of MOSFETs UsingAccumulated Charge Sinks (ACS)—Overview

As described below in more detail, the present disclosure describesmethods and apparatuses for improving semiconductor device linearity(e.g., reducing adverse harmonic distortion and IMD effects) in SOIMOSFETs. In one exemplary embodiment, the method and apparatus improvesthe linearity and controls the harmonic distortion and IMD effects ofthe MOSFET devices by reducing the accumulated charge in the bodies ofthe MOSFET devices. In one embodiment, the present method and apparatusreduces or otherwise controls the accumulated charge in the MOSFETbodies using an accumulated charge sink (ACS) that is operativelycoupled to the MOSFET body. In one embodiment, the present method andapparatus entirely removes all of the accumulated charge from the bodiesof the MOSFET devices. In one described embodiment, the MOSFET is biasedto operate in an accumulated charge regime, and the ACS is used toentirely remove, reduce, or otherwise control, the accumulated chargeand thereby reduce harmonic distortions and IMD that would otherwiseresult. Linearity is also improved in some embodiments by removing orotherwise controlling the accumulated charge thereby improving thefloating body MOSFET BVDSS characteristics.

As noted in the background section above, persons skilled in theelectronic device design and manufacture arts shall appreciate that theteachings herein apply equally to MOSFETs fabricated onSemiconductor-On-Insulator (“SOT”) and Semiconductor-On-Sapphire (“SOS”)substrates. The present teachings can be used in the implementation ofMOSFETs using any convenient semiconductor-on-insulator technology. Forexample, the inventive MOSFETs described herein can be implemented usingcompound semiconductors fabricated on insulating substrates, such asGaAs MOSFETs. As noted above, the present method and apparatus may alsobe applied to silicon-germanium (SiGe) SOT MOSFETs. For simplicity, theembodiments and examples presented herein for illustrative purposesinclude only NMOSFETs, unless otherwise noted. By making well knownchanges to dopants, charge carriers, polarity of bias voltages, etc.,persons skilled in the electronic device design arts will easilyunderstand how these embodiments and examples may be adapted for usewith PMOSFETs.

As noted above, the present disclosure is particularly applicable toFETs and associated applications benefiting from a fully depletedchannel when the FET is operated in the off-state, wherein anaccumulated charge may result. The disclosed method and apparatus foruse in improving the linearity of MOSFETs also finds applicability foruse with partially depleted channels. As known to those skilled in theart, the doping and dimensions of the body vary widely. In an exemplaryembodiment, the body comprises silicon having a thickness ofapproximately 100 angstroms to approximately 2,000 angstroms. In afurther exemplary embodiment, dopant concentration within the FET bodiesranges from no more than that associated with intrinsic silicon toapproximately 1×10¹⁸ active dopant atoms per cm³, resulting infully-depleted transistor operation. In a further exemplary embodiment,dopant concentration within the FET bodies ranges from 1×10¹⁸ to 1×10¹⁹active dopant atoms per cm³ and/or the silicon comprising the bodyranges from a thickness of 2000 angstroms to many micrometers, resultingin partially-depleted transistor operation. As will be appreciated bythose skilled in the electronic design and manufacturing arts, thepresent disclosed method and apparatus for use in improving linearity ofMOSFETs can be used in MOSFETs implemented in a wide variety of dopantconcentrations and body dimensions. The present disclosed method andapparatus therefore is not limited for use in MOSFETs implemented usingthe exemplary dopant concentrations and body dimensions as set forthabove.

According to one aspect of the present disclosure, accumulated chargewithin a FET body is reduced using control methodologies and associatedcircuitry. In one embodiment all of the accumulated charge is removedfrom the FET body. In other embodiments, the accumulated charge isreduced or otherwise controlled. In one embodiment, holes are removedfrom the FET body, whereas in another embodiment, electrons are removedfrom the FET body, as described below in more detail. By removing holes(or electrons) from the FET body using the novel and nonobviousteachings of the present disclosure, voltage induced variations in theparasitic capacitances of the off-state FETs are reduced or eliminated,thereby reducing or eliminating nonlinear behavior of the off-stateFETs. In addition, as described above with reference to FIG. 2A, becausethe body impedance is greatly increased when the accumulated charge isreduced or controlled, there is a beneficial overall reduction in themagnitude of the FET off-state capacitances. Also, as described above,removing or otherwise controlling the accumulated charge in floatingbody MOSFETs improves the BVDSS characteristics of the FET and therebyimproves the linearity of the floating body MOSFET.

Accumulated charge control not only facilitates a beneficial overallreduction in the FET off-state capacitance C_(off) (as described abovewith reference to FIG. 2A and below with reference to FIG. 4H), it alsofacilitates a reduction in C_(off) variations that can occur over timein the presence of a time varying V_(ds) bias voltage. Thus, a reductionof undesirable harmonics generation and intermodulation distortion in RFswitch circuits is obtained using SOI MOSFETs made in accordance withthe present disclosure. Improved SOI MOSFET power handling, linearity,and performance are achieved by devices made in accordance with thepresent teachings. While the methods and apparatuses of the presentdisclosure are capable of fully removing accumulated charge from the FETbodies, those skilled in the electronic device design arts shallappreciate that any reduction of accumulated charge is beneficial.

Reductions in harmonics and intermodulation distortion are generallybeneficial in any semiconductor system, either bulk semiconductor orsemiconductor-on-insulator (SOI) systems. SOI systems include anysemiconductor architecture employing semiconductor-containing regionspositioned above an underlying insulating substrate. While any suitableinsulating substrate can be used in a SOI system, exemplary insulatingsubstrates include silicon dioxide (e.g., a buried oxide layer supportedby a silicon substrate, such as that known as Separation by Implantationof Oxygen (SIMOX)), bonded wafer (thick oxide), glass, and sapphire. Asnoted above, in addition to the commonly used silicon-based systems,some embodiments of the present disclosure may be implemented usingsilicon-germanium (SiGe), wherein the SiGe is used equivalently in placeof Si.

A wide variety of ACS implementations and structures can be used topractice the present disclosed method and apparatus. In accordance withone embodiment of the present method and apparatus, an ACS is used toremove or otherwise control accumulated charge (referenced as 120 inFIG. 1 described above) from the MOSFETs when the MOSFETs are configuredto operate in the accumulated charge regime. By adapting the SOI (orSOS) MOSFETs in accordance with the present teachings, improvedAccumulated Charge Control (ACC) MOSFETs are realized. The ACC MOSFETsare useful in improving performance of many circuits, including RFswitching circuits. Various characteristics and possible configurationsof the exemplary ACC MOSFETs are described in detail below withreference to FIGS. 3A-3K. This section also describes how the exemplaryACS implementations of the present disclosure differ from the bodycontacts of the prior art.

The ACC MOSFET is shown schematically embodied as a four-terminal devicein FIG. 4A. FIGS. 4B-4G show various exemplary simple circuitconfigurations that can be used in removing the accumulated charge fromthe ACC MOSFET when it operates in an accumulated charge regime. Theoperation of the simplified circuit configurations is described in moredetail below with reference to FIGS. 4A-4G. The improvement in off-statecapacitance C_(off) of the ACC MOSFETs, as compared with the off-statecapacitance of the prior art SOI MOSFETs, is described below withreference to FIG. 4H.

The operation of various exemplary RF switch circuits implemented usingthe ACC MOSFETs of the present disclosure is described below withreference to the circuit schematics of FIGS. 5B-5D. Further, anexemplary RF switch circuit using stacked ACC MOSFETs (for increasedpower handling) of the present disclosure is described below withreference to FIG. 6. An exemplary method of improving the linearity ofan SOI MOSFET using an accumulated charge sink (ACS) is described withreference to FIG. 7. Finally, exemplary fabrication methods that may beused to manufacture the ACC MOSFET are described. The various exemplaryACS implementations and structures that can be used to practice thedisclosed method and apparatus are now described with reference to FIGS.3A-3K.

Controlling Accumulated Charge Using an Accumulated Charge Sink (ACS)

FIGS. 3A and 3B are simplified schematic diagrams of a top view of anAccumulated Charge Control (ACC) SOI NMOSFET 300 adapted to controlaccumulated charge 120 (FIG. 1) in accordance with the presentdisclosure. In the exemplary embodiment, a gate contact 301 is coupledto a first end of a gate 302. A gate oxide (not shown in FIG. 3A butshown in FIG. 1) and a body 312 (shown in FIG. 3B) are positioned underthe gate 302. In the exemplary NMOSFET 300 shown, a source 304 and adrain 306 comprise N+ regions. In the exemplary embodiment, the ACCNMOSFET 300 includes an accumulated charge sink (ACS) 308 comprising aP− region. The ACS 308 is coupled to and is in electrical communicationwith the body 312 which also comprises a P− region. An electricalcontact region 310 provides electrical connection to the ACS 308. Insome embodiments, the electrical contact region 310 comprises a P+region. As shown in FIG. 3A, the electrical contact region 310 iscoupled to and is in electrical communication with the ACS 308.

Those skilled in the arts of electronic devices shall understand thatthe electrical contact region 310 may be used to facilitate electricalcoupling to the ACS 308 because in some embodiments it may be difficultto make a direct contact to a lightly doped region. In addition, in someembodiments the ACS 308 and the electrical contact region 310 may becoextensive. In another embodiment, the electrical contact region 310comprises an N+ region. In this embodiment, the electrical contactregion 310 functions as a diode connection to the ACS 308, whichprevents positive current flow into the ACS 308 (and also preventspositive current flow into the body 312) under particular biasconditions, as described below in more detail.

FIG. 3B is an alternative top view of the ACC SOI NMOSFET 300 of FIG.3A, illustrating the ACC NMOSFET 300 without its gate contact 301, gate302, and gate oxide being visible. This view allows the body 312 to bevisible. FIG. 3B shows the coupling of the ACS 308 to one end of thebody 312. In one embodiment, the body 312 and the ACS 308 comprise acombined P− region that may be produced by a single ion-implantationstep. In another embodiment, the body 312 and ACS 308 comprise separateP− regions that are coupled together.

As is well known to those skilled in the electronic device design arts,in other embodiments, the ACC NMOSFET 300 of FIGS. 3A and 3B can beimplemented as an ACC PMOSFET simply by reversing the dopant materialsused to implement the various FET component regions (i.e., replacep-type dopant material with n-type dopant material, and vice versa).More specifically, in an ACC PMOSFET, the source and drain comprise P+regions, and the body comprises an N− region. In this embodiment, theACS 308 also comprises an N− region. In some embodiments of the ACCPMOSFET, the electrical contact region 310 may comprise an N+ region. Inother embodiments of the ACC PMOSFETs, the region 310 comprises a P+region, which functions as a diode connection to the ACS 308 and therebyprevents current flow into the ACS 308 under particular bias conditions.

Prior Art Body Contacts Distinguished from the Disclosed ACS

According to the present disclosure, the ACS 308 used to implement ACCSOT MOSFETs includes novel features in structure, function, operationand design that distinguish it from the so-called “body contacts” (alsosometimes referred to as “body ties”, usually when the “body contact” isdirectly connected to the source) that are well known in the prior art.

Exemplary references relating to body contacts used in prior art SOTMOSFETs include the following: (1) F. Hameau and O. Rozeau,Radio-Frequency Circuits Integration Using CMOS SOT 0.25 μm Technology,”2002 RF IC Design Workshop Europe, 19-22 Mar. 2002, Grenoble, France;(2) J. R. Cricci et al., “Silicon on Sapphire MOS Transistor,” U.S. Pat.No. 4,053,916, Oct. 11, 1977; (3) O. Rozeau et al., “SOT TechnologiesOverview for Low-Power Low-Voltage Radio-Frequency Applications,” AnalogIntegrated Circuits and Signal Processing, 25, pp. 93-114, Boston,Mass., Kluwer Academic Publishers, November 2000; (4) C. Tinella et al.,“A High-Performance CMOS-SOT Antenna Switch for the 2.5-5-GHz Band,“IEEE Journal of Solid-State Circuits, Vol. 38, No. 7, July, 2003; (5)H. Lee et al., “Analysis of body bias effect with PD-SOI for analog andRF applications,” Solid State Electron., Vol. 46, pp. 1169-1176, 2002;(6) J.-H. Lee, et al., “Effect of Body Structure on Analog Performanceof SOT NMOSFETs,” Proceedings, 1998 IEEE International SOT Conference,5-8 Oct. 1998, pp. 61-62; (7) C. F. Edwards, et al., The Effect of BodyContact Series Resistance on SOT CMOS Amplifier Stages,” IEEETransactions on Electron Devices, Vol. 44, No. 12, December 1997 pp.2290-2294; (8) S. Maeda, et al., Substrate-bias Effect and Source-drainBreakdown Characteristics in Body-tied Short-channel SOT MOSFET's,” IEEETransactions on Electron Devices, Vol. 46, No. 1, January 1999 pp.151-158; (9) F. Assaderaghi, et al., “Dynamic Threshold-voltage MOSFET(DTMOS) for Ultra-low Voltage VLSI,” IEEE Transactions on ElectronDevices, Vol. 44, No. 3, March 1997, pp. 414-422; (10) G. O. Workman andJ. G. Fossum, “A Comparative Analysis of the Dynamic Behavior of BTG/SOIMOSFETs and Circuits with Distributed Body Resistance,” IEEETransactions on Electron Devices, Vol. 45, No. 10, October 1998 pp.2138-2145; and (11) T.-S. Chao, et al., “High-voltage andHigh-temperature Applications of DTMOS with Reverse Schottky Barrier onSubstrate Contacts,” IEEE Electron Device Letters, Vol. 25, No. 2,February 2004, pp. 86-88.

As described herein, applications such as RF switch circuits, may useSOI MOSFETs operated with off-state bias voltages, for which accumulatedcharge may result. The SOI MOSFETs are defined herein as operatingwithin the accumulated charge regime when the MOSFETs are biased in theoff-state, and when carriers having opposite polarity to the channelcarriers are present in the channel regions of the MOSFETs. In someembodiments, the SOI MOSFETs may operate within the accumulated chargeregime when the MOSFETs are partially depleted yet still biased tooperate in the off-state. Significant benefits in improving nonlineareffects on source-drain capacitance can be realized by removing orotherwise controlling the accumulated charge according to the presentteachings. In contrast to the disclosed techniques, none of the citedprior art teach or suggest ACS methods and apparatuses that are uniquelyuseful for removing or controlling accumulated charge. Nor are theyinformed regarding problems caused by the accumulated charge such asnonlinear effects on the off-state source-drain capacitance C_(off).Consequently, the prior art body contacts described in the referencescited above differ greatly (in structure, function, operation anddesign) from the ACSs described with reference to FIGS. 3A-4D.

In one example, the ACS 308 operates effectively to remove or otherwisecontrol the accumulated charge from the SOI NMOSFET 300 using a highimpedance connection to and throughout the body 312. High impedance ACSsmay be used because the accumulated charge 120 is primarily generated byphenomena (e.g., thermal generation) that take a relatively long periodof time to produce significant accumulated charge. For example, atypical time period for producing non-negligible accumulated charge whenthe NMOSFET operates in the accumulated charge regime is approximately afew milliseconds or greater. Such relatively slow generation ofaccumulated charge corresponds to very low currents, typically less than100 nA/mm of transistor width. Such low currents can be effectivelyconveyed even using very high impedance connections to the body.According to one example, the ACS 308 is implemented with a connectionhaving a resistance of greater than 10⁶ ohms. Consequently, the ACS 308is capable of effectively removing or otherwise controlling theaccumulated charge 120 even when implemented with a relatively highimpedance connection, relative to the low impedance prior art bodycontacts.

In stark contrast, the prior art teachings of body contacts described inthe references cited above require low impedance (high efficiency)access to the body regions of SOI MOSFETs for proper operation (see,e.g., references (3), (6), and (7) above). A principal reason for thisrequirement is that the prior art body contacts are primarily directedto reducing the adverse effects on SOI MOSFET functions caused by muchfaster and more effective electron-hole pair generation processes thanoccur when the FET is operated in the accumulated charge regime. Forexample, in some prior art MOSFETs not operated in the accumulatedcharge regime, electron-hole pair carriers are generated as a result ofimpact ionization. Impact ionization produces electron-hole pairs at amuch faster rate than occurs when the FET is operated in the accumulatedcharge regime.

The relative rates for electron-hole pair generation by impactionization versus the pair generation processes causing accumulatedcharge can be estimated from the roll-off frequencies for the twophenomena. For example, reference (3) cited above indicates roll-offfrequencies for impact ionization effects in the range of 10⁵ Hz. Incontrast, a roll-off frequency for the accumulated charge effects hasbeen observed to be in the range of 10³ Hz or less, as indicated byrecovery times for odd harmonics. These observations indicate that theACS 308 can effectively control accumulated charge using an impedancethat is at least 100 times larger than required of prior art bodycontacts used in controlling impact ionization charge, for example.Further, because impact ionization primarily occurs when the SOI MOSFEToperates in an on-state, the effects of impact ionization can beamplified by on-state transistor operation. Low impedance body contactsto and throughout a body region is even more critical in theseenvironments in order to control the effects of impact ionization underthe on-state conditions.

In stark contrast, the ACS 308 of the present teachings removes orotherwise controls the accumulated charge only when the ACC SOI MOSFEToperates in the accumulated charge regime. By definition, the FET is inthe off-state in this regime, so there is no requirement to removeimpact ionization as amplified by an on-state FET. Therefore, a highimpedance ACS 308 is perfectly adequate for removing the accumulatedcharge under these operating conditions. The prior art requirements forlow impedance body connections results in numerous problems ofimplementation that are overcome by the present teachings, as describedbelow in more detail.

In addition, the ACS 308 may be implemented with much lowersource-to-drain parasitic capacitance as compared to the body contactsof the prior art. The above-described low impedance connection to theSOI MOSFET body required of the prior art body contacts necessitatesproximity of the contacts to the entire body. This may require aplurality body contact “fingers” that contact the body at differentlocations along the body. The low impedance connection to the body alsonecessitates proximity of the prior art body contacts to the source anddrain. Because of parasitic capacitances produced by such body contacts,the cited prior art references teach away from the use of suchstructures for many high frequency applications such as RF. In starkcontrast, the ACS 308 of the present disclosure may be positioned aselected distance away from the source 304 and the drain 306, and theACS 308 may also be coupled to the body 312 at a first distal end of thebody 312 (shown in FIGS. 3A and 3B). Arranged in this manner, the ACS308 makes minimal contact (as compared to the prior art body contactsthat may contact the body at many locations along the body) with thebody 312. This configuration of the ACS 308 with the MOSFET eliminatesor greatly reduces the parasitic capacitances caused by a more proximatepositioning of the ACS 308 relative to the source, drain, and body.Further, the ACS 308 may be implemented in SOI MOSFETs operated with adepleted channel. In general, the cited prior art references teach awayfrom the use of body contacts for this environment (see, e.g., reference(3), cited above).

Further, because impact ionization hole currents are much larger (in therange of 5,000 nA per mm body width) than for accumulated chargegeneration (less than approximately 100 nA per mm body width), the priorart does not teach how to effectively implement very large body widths(i.e., much greater than approximately 10 μm). In contrast, the ACS 308of the present disclosed device may be implemented in SOI MOSFETs havingrelatively large body widths. This provides improvements in on-stateconductance and transconductance, insertion loss and fabrication costs,particularly for RF switch devices. According to the prior art teachingscited above, larger body widths adversely affect the efficient operationof body contacts because their impedances are necessarily therebyincreased. Although the cited prior art suggests that a plurality offingers may be used to contact the body at different locations, theplurality of fingers adversely affects parasitic source-to-draincapacitances, as described above.

For these reasons, and for the reasons described below in more detail,the present disclosure provides novel MOSFET devices, circuits andmethods that overcome the limitations according to the prior artteachings as cited above.

FIG. 3C is a cross-sectional perspective schematic of an ACC SOI NMOSFET300′ adapted to control accumulated charge in accordance with thedisclosed method and apparatus. In the example shown in FIG. 3C, the ACCNMOSFET 300′ includes four terminals that provide electrical connectionto the various FET component regions. In one embodiment, the terminalsprovide means for connecting external integrated circuit (IC) elements(such as metal leads, not shown) to the various FET component regions.Three of the terminals shown in FIG. 3C are typically available in priorart FET devices. For example, as shown in FIG. 3C, the ACC NMOSFET 300′includes a gate terminal 302′ that provides electrical connection to thegate 302. Similarly, the ACC NMOSFET 300′ includes source and drainterminals 304′, 306′ that provide electrical connection to the source304 and drain 306, respectively. As is well known in the electronicdesign arts, the terminals are coupled to their respective FET componentregions (i.e., gate, drain and source) via so-called “ohmic” (i.e., lowresistance) contact regions. The manufacturing and structural detailsassociated with the coupling of the various FET terminals to the FETcomponent regions are well known in the art, and therefore are notdescribed in more detail here.

As described above with reference to FIGS. 3A and 3B, the ACC NMOSFET300′ is adapted to control accumulated charge when the NMOSFET operatesin the accumulated charge regime. To this end, in the exemplaryembodiment shown in FIG. 3C, the ACC NMOSFET 300′ includes a fourthterminal that provides electrical connection to the body 312, andthereby facilitates reduction (or other control) of the accumulatedcharge when the FET 300′ operates in the accumulated charge regime. Morespecifically, and referring again to FIG. 3C, the ACC NMOSFET includes a“body” terminal, or Accumulated Charge Sink (ACS) terminal 308′. The ACSterminal 308′ provides an electrical connection to the ACS 308 (notshown in FIG. 3C, but shown in FIGS. 3A and 3B) and to the body 312.Although the ACS terminal 308′ is shown in FIG. 3C as being physicallycoupled to the body 312, those skilled in the electronic design artsshall understand that this depiction is for illustrative purposes only.The direct coupling of the ACS terminal 308′ to the body 312 shown inFIG. 3C illustrates the electrical connectivity (i.e., not the physicalcoupling) of the terminal 308′ with the body 312. Similarly, the otherterminals (i.e., terminals 302′, 304′ and 306′) are also shown in FIG.3C as being physically coupled to their respective FET componentregions. These depictions are also for illustrative purposes only.

In most embodiments, as described above with reference to FIGS. 3A-3B,and described further below with reference to FIGS. 3D-3K, the ACSterminal 308′ provides the electrical connection to the body 312 viacoupling to the ACS 308 via the electrical contact region 310. However,the present disclosure also contemplates embodiments where the couplingof the ACS terminal 308′ is made directly to the body 312 (i.e., nointermediate regions exist between the ACS terminal 308′ and the body312).

In accordance with the disclosed method and apparatus, when the ACCNMOSFET 300′ is biased to operate in the accumulated charge regime(i.e., when the ACC NMOSFET 300′ is in the off-state, and there is anaccumulated charge 120 of P polarity (i.e., holes) present in thechannel region of the body 312), the accumulated charge is removed orotherwise controlled via the ACS terminal 308′. When accumulated charge120 is present in the body 312, the charge 312 can be removed orotherwise controlled by applying a bias voltage (Vb (for “body”) orV_(ACS) (ACS bias voltage)) to the ACS terminal 308′. In general, theACS bias voltage V_(ACS) applied to the ACS terminal 308′ may beselected to be equal to or more negative than the lesser of the sourcebias voltage Vs and drain bias voltage Vd. More specifically, in someembodiments, the ACS terminal 308′ can be coupled to various accumulatedcharge sinking mechanisms that remove (or “sink”) the accumulated chargewhen the FET operates in the accumulated charge regime. Severalexemplary accumulated charge sinking mechanisms and circuitconfigurations are described below with reference to FIGS. 4A-5D.

Similar to the prior art NMOSFET 100 described above with reference toFIG. 1, the ACC SOI NMOSFET 300′ of FIG. 3C can be biased to operate inthe accumulated charge regime by applying specific bias voltages to thevarious terminals 302′, 304′, and 306′. In one exemplary embodiment, thesource and drain bias voltages (Vs and Vd, respectively) are zero (i.e.,the terminals 304′ and 306′ are connected to ground). In this example,if the gate bias voltage (Vg) applied to the gate terminal 302′ issufficiently negative with respect to the source and drain biasvoltages, and with respect to V_(th) (for example, if V_(th) isapproximately zero, and if Vg is more negative than approximately −1 V),the ACC NMOSFET 300′ operates in the off-state. If the ACC NMOSFET 300′continues to be biased in the off-state, the accumulated charge (holes)will accumulate in the body 312. Advantageously, the accumulated chargecan be removed from the body 312 via the ACS terminal 308′. In someembodiments, as described below in more detail with reference to FIG.4B, the ACS terminal 308′ is coupled to the gate terminal 302′ (therebyensuring that the same bias voltages are applied to both the gate (Vg)and the body (shown in FIG. 3C as “Vb” or “V_(ACS)”).

However, those skilled in the electronics design arts shall appreciatethat a myriad of bias voltages can be applied to the four deviceterminals while still employing the techniques of the present disclosedmethod and apparatus. As long as the ACC SOI NMOSFET 300′ is biased tooperate in the accumulated charge regime, the accumulated charge can beremoved or otherwise controlled by applying a bias voltage V_(ACS) tothe ACS terminal 308′, and thereby remove the accumulated charge fromthe body 312.

For example, in one embodiment wherein the ACC NMOSFET 300′ comprises adepletion mode device, V_(th) is negative by definition. In thisembodiment if both the Vs and Vd bias voltages comprise zero volts(i.e., both terminals tied to circuit ground node), and a gate bias Vgapplied to the gate terminal 302′ is sufficiently negative to V_(th)(for example, Vg is more negative than approximately −1 V relative toV_(th)), holes may accumulate under the gate oxide 110 thereby becomingthe accumulated charge 120. In this example, in order to remove theaccumulated holes (i.e., the accumulated charge 120) from the FET body312, the voltage V_(ACS) applied to the ACS 308 may be selected to beequal to or more negative than the lesser of Vs and Vd.

In other examples, the source and drain bias voltages, Vs and Vd,respectively, may comprise voltage other than zero volts. According tothese embodiments, the gate bias voltage Vg must be sufficientlynegative to both Vs and Vd (in order for Vg to be sufficiently negativeto V_(th), for example) in order to bias the NMOSFET in the off-state.As described above, if the NMOSFET is biased in the off-state for asufficiently long time period (approximately 1-2 ms, for example) anaccumulated charge will accumulate under the gate oxide. In theseembodiments, as noted above, in order to remove the accumulated charge120 from the body 312, the ACS bias voltage V_(ACS) applied to the ACSterminal 308′ may be selected to be equal to or more negative than thelesser of Vs and Vd.

It should be noted that, in contrast to the examples described above,the prior art body contacts are implemented largely for purposes ofmitigating the adverse effects caused by impact ionization.Consequently, the prior art body contacts are typically tied to thesource of the MOSFET. In order to effectively control, reduce, orentirely remove the accumulated charge in an NMOSFET, V_(ACS) should, inthe exemplary embodiments, be equal to or more negative than the lesserof Vs and Vd. Those skilled in the electronic device design arts shallappreciate that different Vs, Vd, Vg and V_(ACS) bias voltages may beused when the ACC MOSFET comprises a PMOSFET device. Because the priorart body contacts are typically tied to the source, this implementationcannot be effected using the prior art body contact approach.

FIG. 3D is a simplified schematic diagram of a top view of an ACC SOINMOSFET 300″ adapted to control accumulated charge 120 (FIG. 1) inaccordance with the present disclosure. FIG. 3D shows the ACC NMOSFET300″ without its gate contact 301, gate 302, and gate oxide beingvisible. The ACC NMOSFET 300″ of FIG. 3D is very similar in design tothe ACC NMOSFET 300 described above with reference to FIGS. 3A and 3B.For example, similar to the ACC NMOSFET 300, the ACC NMOSFET 300″includes a source 304 and drain 306 comprising N+ regions. The ACCNMOSFET 300″ also includes an accumulated charge sink (ACS) 308comprising a P− region. As shown in FIG. 3D, the P− region thatcomprises the ACS 308 abuts (i.e., is directly adjacent) the body 312,which also comprises a P− region. Similar to the ACC NMOSFET 300, theACC NMOSFET 300″ includes an electrical contact region 310 that provideselectrical connection to the ACS 308. As noted above, in someembodiments, the electrical contact region 310 comprises a P+ region. Inanother embodiment, the electrical contact region 310 may comprise an N+region (which thereby prevents positive current flow into the body 312as noted above). As shown in FIG. 3D, the electrical contact region 310is formed in the ACC NMOSFET 300″ directly adjacent the ACS 308. The ACCSOI NMOSFET 300″ functions to control accumulated charge similarly tothe operation of the ACC NMOSFETs described above with reference toFIGS. 3A-3C.

FIG. 3E is a simplified schematic diagram of a top view of an ACC SOINMOSFET 300′″ adapted to control accumulated charge in accordance withthe present disclosure. The ACC NMOSFET 300′″ is very similar in designand function to the ACC NMOSFETs described above with reference to FIGS.3A-3D. FIG. 3E shows a dashed cross-sectional view line A-A′ taken alongthe approximate center of the NMOSFET 300′″. This cross-sectional viewis used herein to describe structural and performance characteristics ofsome exemplary prior art MOSFETS and some embodiments of the ACC NMOSFETthat may occur as a result of the fabrication processes. Details of thiscross-sectional view A-A′ are now described with reference to FIG. 3F.

View line A-A′ slices through the following component regions of the ACCNMOSFET 300′″: the P+ electrical contact region 310, the ACS 308 (shownin FIG. 3E, but not shown in FIG. 3F), a P+ overlap region 310′, a gateoxide 110, and a poly-silicon gate 302. In some embodiments, during thefabrication process, when the region 310 is doped with p-type dopantmaterial, proximate the P− body region, some additional P+ doping may beimplanted (i.e., the p-type dopant material may overlap) into the P+overlap region 310′ of the poly-silicon gate 302. In some embodiments,such overlapping is performed intentionally to ensure that all of thegate oxide 110 is completely covered by the P+ region (i.e., to ensurethat no gap exists on the edge of the oxide 110 between the gate 302 andthe P+ region 310). This, in turn, aids in providing a minimum impedanceconnection between the P+ region 310 and the body 312.

Although the present teachings encompass such embodiments describedabove, those skilled in the electronic device design and manufacturingarts shall recognize that such low-resistance connections are notrequired. Therefore, disadvantages associated with the embodiment shownin FIG. 3H, as described below in more detail, can be overcome by usingother embodiments described herein (for example, the embodiments 300 and300″ described below with reference to FIGS. 3G and 3J, respectively),in which gaps are intentionally implemented between the P+ region 310and the body 312. In one exemplary embodiment, the P+ overlap region310′ overlaps the oxide 110 by approximately 0.2-0.7 microns. Thoseskilled in the MOSFET design and manufacturing arts shall appreciatethat other overlap region dimensions can be used in practicing thepresent disclosed method and apparatus. In some embodiments, as shown inFIG. 3F, for example, the remaining area over the gate oxide 110 andover the P− body is doped with n-type dopant material (i.e., itcomprises an N+ region).

Referring again to FIG. 3F, owing to the presence of the P+ overlapregion 310′ over the gate oxide 110, over the body 312, and proximate anedge 340 of the poly-silicon gate 302, an increased threshold voltageregion is created in the NMOSFET 300′″. More specifically, due to the P+doping (in the P+ overlap region 310′) proximate the edge 340 of thegate 302 over the channel region of the body 312, a region of increasedthreshold voltage is formed in that region of the MOSFET 300′″. Theeffects of the region of increased threshold voltage are now describedin more detail with reference to FIGS. 3H and 3I.

FIG. 3I shows a plot 380 of inversion channel charge versus applied gatevoltage for an ACC NMOSFET. The plot 380 shown in FIG. 3I illustratesone effect of the above-described increased threshold voltage that canoccur in prior art MOSFETs, and in some embodiments of the present ACCNMOSFETs due to certain manufacturing processes. As described in moredetail below, the increased threshold voltage region, shown in FIG. 3Hand described in more detail below, also occurs in prior art MOSFETdesigns due to the proximity of body ties to the FET body. As describedbelow in more detail with reference to FIG. 3J, for example, the presentdisclosed method and apparatus can be used to reduce or eliminate theregion of increased threshold voltage found in some prior art SOI MOSFETdesigns.

FIG. 3H shows one embodiment of an ACC NMOSFET without its gate contact,gate, and gate oxide being visible. The MOSFET region of increasedthreshold voltage described above with reference to FIGS. 3E and 3F isshown in FIG. 3H as occurring in the region encompassed by the ellipse307. As will be well understood by those skilled in the electronicdesign and manufacturing arts, for the reasons set forth above withreference to FIGS. 3E and 3F, due to the increased threshold voltage,the region 307 of the ACC MOSFET shown in FIG. 3H effectively “turns on”after the rest of the ACC MOSFET channel region.

The increased threshold voltage can be reduced by reducing the size ofthe region 307. Eliminating the region 307 altogether eliminates thethreshold voltage increase. Because the threshold voltage increase canincrease harmonic and intermodulation distortion of the “on” stateMOSFET, eliminating this effect improves MOSFET performance. Theincreased threshold voltage also has the detrimental effect ofincreasing the MOSFET on-resistance (i.e., the resistance presented bythe MOSFET when it is in the on-state (conducting state), whichdetrimentally impacts the MOSFET insertion loss.

In one exemplary embodiment, as shown, for example in the embodiments ofthe ACC NMOSFET 300 described above with reference to FIGS. 3A and 3B,and as described below in more detail with reference to thecross-sectional view of the ACC MOSFET 300 of FIG. 3G, the detrimentaleffects associated with threshold voltage increase are mitigated orovercome by positioning the P+ region 310 a selected distance away froman edge of the poly-silicon gate 302. This approach is shown both in thetop view of the ACC MOSFET 300 of FIG. 3A, and in the cross-sectionalview of the ACC MOSFET 300 shown in FIG. 3G. As shown in thecross-sectional view of the ACC MOSFET 300 of FIG. 3G, the P+ region 310does not extend all the way to the edge 340 of the poly-silicon gate302. This is in stark contrast to the embodiment 300′″ shown in FIG. 3F,where the P+ region 310′ extends all the way to the gate edge 340. Bypositioning the P+ region 310 a distance away from the gate edge 340 asshown in the embodiment 300 of FIG. 3G, no P+ region is positionedproximate the poly-silicon gate 302 (i.e., there is no P+ region presentin the poly-silicon gate 302).

This configuration of the P+ region 310 eliminates or greatly reducesthe problems associated with threshold voltage increase as describedabove. As described above with reference to FIGS. 3A and 3B, and withreference to the comparisons to the prior art body contact references,the relatively high impedance of the ACS 308 P− region (shown in FIG.3A) between the P+ region 310 and the gate 302 does not adversely affectthe performance of the ACC NMOSFET 300. As described above, theaccumulated charge can be effectively removed even using a relativelyhigh impedance ACS connection.

In another exemplary embodiment, as described below with reference toFIG. 3J, the threshold voltage increase is removed by positioning the P+region 310 (and the ACS 308) a distance away from the body 312. Becausethe electrical connectivity between the ACS 308 and the body 312 hasrelatively high impedance when the small region of P+ 310 is positioneda distance away from the body 312, this approach is never taught orsuggested by the body contact prior art references (which require lowimpedance contacts as described above). This improved embodiment isdescribed next with reference to FIG. 3J.

FIG. 3J is a simplified top view schematic of another embodiment of anACC SOI NMOSFET 300″″ adapted to control accumulated charge andconfigured in a “T-gate” configuration. FIG. 3J shows the ACC NMOSFET300″″ without its gate contact 301, gate 302, and gate oxide beingvisible. The gate (not shown in FIG. 3J) and the body 312 are configuredas “supporting” members of the “T-gate” configured ACC MOSFET 300″″(i.e., they comprise the “bottom” portion of the “T-shaped” FET). These“supporting” members “support” the “supported” member of the T-gateconfigured MOSFET 300″, which comprises the ACS 308 as shown in FIG. 3J(i.e., the ACS 308 comprises the “top” portion of the “T-shaped” FET).As shown in FIG. 3J, the ACC NMOSFET 300″″ includes a small P+ region310 conjoined to an ACS 308. As shown in FIG. 3J, the P+ region 310 (andthus the ACS external electrical connection) is disposed a selecteddistance away from the body 312. The total impedance of the electricalconnection from the body 312, through the ACS 308, and to the P+ region310 is increased by positioning the P+ region 310 a selected distanceaway from the body 312. However, as described above, the present ACCNMOSFET 300″″ works perfectly well to remove accumulated charge evenusing relatively high impedance ACS connections. For the reasonsdescribed above with reference to FIGS. 3A and 3B, due to the nature ofthe accumulated charge when the NMOSFET 300″″ operates in theaccumulated charge regime, the ACC NMOSFET 300″″ does not require lowimpedance ACS electrical connections in order to remove accumulatedcharge from the body 312. Rather, an ACS connection of relatively largeimpedance may be used in practicing the present teachings, withcorresponding improvements in NMOSFET performance as described above(e.g., reductions in parasitic capacitance as compared with prior artlow impedance body contacts). However, in other embodiments, if desired,a low impedance ACS connection may be used to practice the disclosedmethod and apparatus for use in improving linearity characteristics ofSOI MOSFETs.

Moreover, as described above with reference to FIG. 3H, the embodimentof FIG. 3J improves device performance owing to the fact that the smallP+ region 310 is positioned a distance away from the body 312. Becausethe small P+ region 310 is positioned a distance away from the body 312,the threshold voltage increase is reduced or entirely eliminated,together with the consequent adverse performance effects describedabove.

FIG. 3K is a simplified top view schematic of another embodiment of anACC SOI NMOSFET 300′″″ adapted to control accumulated charge andconfigured in an “H-gate” configuration. FIG. 3K shows the ACC NMOSFET300′″″ without its gate contact 301, gate 302, and gate oxide beingvisible. With the exception of some structural differences describedherein, the ACC NMOSFET 300′″″ is very similar in design and function tothe ACC NMOSFETs described above with reference to FIGS. 3A-3D and 3J.As shown in FIG. 3K, the ACC NMOSFET 300′″ includes two ACSs, 308 and308″, disposed at opposite ends of the H-gate ACC NMOSFET 300′. P+regions 310 and 310″ are formed to abut their respective ACSs, 308 and308″, and provide electrical contact thereto. In accordance with thedisclosed method and apparatus, as described above, when the ACC NMOSFET300′″ is biased to operate in the accumulated charge regime, theaccumulated charge is removed or otherwise controlled via the two ACSs308 and 308″.

It shall be understood by those skilled in the electronic device designarts that although the illustrated embodiment shows the ACSs 308 and308″ extending approximately the entire width of the ACC NMOSFET 300′″,the ACSs 308 and 308″ may also comprise much narrower (or wider)regions, and still function perfectly well to remove or otherwisecontrol the accumulated charge. Also, in some embodiments, it is notnecessary that the impedance of the ACS 308 matches the impedance of theACS 308″. It will further be understood by the skilled person that theACSs 308 and 308″ may comprise different sizes and configurations (i.e.,rectangular, square, or any other convenient shape), and may also bepositioned at various distances away from the body 312 (i.e., notnecessarily the same distance away from the body 312). As describedabove with reference to FIG. 3J, when the ACS 308 is positioned aselected distance away from the body 312, the problems associated withthreshold voltage increase are reduced or eliminated.

Four-Terminal ACC MOSFET Devices—Simple Circuit Configurations

The SOI NMOSFET 300 of FIGS. 3A and 3B may be implemented as a fourterminal device, as illustrated schematically in FIG. 4A. As shown inthe improved ACC SOI NMOSFET 300 of FIG. 4A, a gate terminal 402 iselectrically coupled to the gate contact 301 (e.g., FIG. 3A) and isanalogous to the gate terminal 302′ shown in FIG. 3C. The gate contact301 is electrically coupled to the gate 302 (e.g., FIGS. 3A and 3C).Similarly, a source terminal 404 is electrically coupled to the source304 (e.g., FIGS. 3A-3C) and is analogous to the source terminal 304′ ofFIG. 3C. Similarly, a drain terminal 406 is electrically coupled to thedrain 306 (e.g., FIGS. 3A-3C) and is analogous to the drain terminal306′ of FIG. 3C. Finally, the ACC NMOSFET 300 includes an ACS terminal408 that is electrically coupled to the ACS 308 (e.g., see FIGS. 3A-3B,and FIGS. 3D, 3J-3K) via the region 310. Those skilled in the electronicdesign and manufacturing arts shall understand that the region 310 maybe used in some embodiments to facilitate electrical coupling to the ACS308 because, in some embodiments, it may be difficult to make a directcontact to a lightly doped region (i.e., the ACS 308). The ACS terminal408 is analogous to the ACS terminal 308′ shown in FIG. 3C.

The ACC SOI NMOSFET 300 of FIG. 4A may be operated using varioustechniques and implemented in various circuits in order to controlaccumulated charge present in the FET when it is operating in anaccumulated charge regime. For example, in one exemplary embodiment asshown in FIG. 4B, the gate and ACS terminals, 402 and 408, respectively,are electrically coupled together. In one embodiment of the simplifiedcircuit shown in FIG. 4B, the source and drain bias voltages applied tothe terminals 404 and 406, respectively, may be zero. If the gate biasvoltage (Vg) applied to the gate terminal 402 is sufficiently negativewith respect to the source and drain bias voltages applied to theterminals 404 and 406, and with respect to the threshold voltage V_(th),(for example, if V_(th) is approximately zero, and if Vg is morenegative than approximately −1 V) the ACC NMOSFET 300 operates in theaccumulated charge regime. As described above with reference to FIG. 3C,for example, when the MOSFET operates in this regime, accumulated charge(holes) may accumulate in the body of the NMOSFET 300.

Advantageously, the accumulated charge can be removed via the ACSterminal 408 by connecting the ACS terminal 408 to the gate terminal 402as shown. This configuration ensures that when the FET 300 is in theoff-state, it is held in the correct bias region to effectively removeor otherwise control the accumulated charge. As shown in FIG. 4B,connecting the ACS terminal 408 to the gate ensures that the same biasvoltages are applied to both the gate (Vg) and the body (shown in FIG.3C as “Vb” or “V_(ACS)”). Because the bias voltage V_(ACS) is the sameas the gate voltage Vg in this embodiment, the accumulated charge is nolonger trapped below the gate oxide (by attraction to the gate bias Vg)because it is conveyed to the gate terminal 402 via the ACS terminal408. The accumulated charge is thereby removed from the body via the ACSterminal 408.

In other exemplary embodiments, as described above with reference toFIG. 3C, for example, Vs and Vd may comprise nonzero bias voltages.According to these examples, Vg must be sufficiently negative to both Vsand Vd in order for Vg to be sufficiently negative to V_(th) to turn theNMOSFET 300 off (i.e., operate the NMOSFET 300 in the off-state). Whenso biased, as described above, the NMOSFET 300 may enter the accumulatedcharge regime and thereby have accumulated charge present in the body.For this example, the voltage V_(ACS) may also be selected to be equalto Vg by connecting the ACS terminal 408 to the gate terminal 402,thereby conveying the accumulated charge from the body of the ACCNMOSFET, as described above.

In another exemplary embodiment, as described above, the ACC NMOSFET 300comprises a depletion mode device. In this embodiment, the thresholdvoltage, V_(th) is, by definition, less than zero. For Vs and Vd both atzero volts, when a gate bias Vg sufficiently negative to V_(th) isapplied to the gate terminal 402 (for example, Vg more negative thanapproximately −1 V relative to V_(th)), holes may accumulate under thegate oxide and thereby comprise an accumulated charge. For this example,the voltage V_(ACS) may also be selected to be equal to Vg by connectingthe ACS terminal 408 to the gate terminal 402, thereby conveying theaccumulated charge from the ACC NMOSFET as described above.

In some embodiments of the improved ACC SOI NMOSFET 300, such as thatdescribed above with reference to FIG. 4B, when the FET is biased on,diodes formed at the edge of the device (such as described above withreference to the interface between the ACS 308 and the drain 304 (andthe source 306) as shown in FIG. 3D) may become forward biased therebyallowing current to flow into the source and drain regions. In additionto wasting power, this may introduce nonlinearity into the NMOSFET. Thenonlinearity results because the current that flows as a result of theforward biased interface diodes comprises nonlinear current. As Vgs andVgd are reduced in that region of the device, the on resistance Ron atthe edge of the device is increased. As is well known, and for thereasons set forth above, if the interface diodes formed at the edge ofthe device become forward biased, the device on-state characteristicsare consequently dramatically adversely affected. Those skilled in theelectronic device design arts shall understand that the configurationshown in FIG. 4B limits application of a gate bias voltage Vgs toapproximately 0.7 Volts. The simplified circuit shown in FIG. 4C can beused to overcome these problems.

Another exemplary simplified circuit using the improved ACC SOI NMOSFET300 is shown in FIG. 4C. As shown in FIG. 4C, in this embodiment, theACS terminal 408 may be electrically coupled to a diode 410, and thediode 410 may, in turn, be coupled to the gate terminal 402. Thisembodiment may be used to prevent a positive current flow into theMOSFET body 312 caused by a positive Vg-to-Vs (or, equivalently, Vgs,where Vgs=Vg−Vs) bias voltage, as may occur, for example, when the SOINMOSFET 300 is biased into an on-state condition.

As with the device shown in FIG. 4B, when biased off, the ACS terminalvoltage V_(ACS) comprises the gate voltage plus a voltage drop acrossthe diode 410. At very low ACS terminal current levels, the voltage dropacross the diode 410 typically also is very low (e.g., <<500 mV, forexample, for a typical threshold diode). The voltage drop across thediode 410 can be reduced to approximately zero by using other diodes,such as a 0Vf diode, for example. In one embodiment, reducing thevoltage drop across the diode is achieved by increasing the diode 410width. Additionally, maintaining the ACS-to-source or ACS-to-drainvoltage (whichever bias voltage of the two bias voltages is lower)increasingly negative, also improves the linearity of the ACC MOSFETdevice 300.

When the SOI NMOSFET 300 is biased in an on condition, the diode 410 isreverse-biased, thereby preventing the flow of positive current into thesource and drain regions. The reverse-biased configuration reduces powerconsumption and improves linearity of the device. The circuit shown inFIG. 4C therefore works well to remove accumulated charge from the ACCMOSFET body when the FET is in the off-state and is operated in theaccumulated charge regime. It also permits almost any positive voltageto be applied to the gate voltage Vg. This, in turn, allows the ACCMOSFET to effectively remove accumulated charge when the device operatesin the off-state, yet assume the characteristics of a floating bodydevice when the device operates in the on-state.

With the exception of the diode 410 used to prevent the flow of positivecurrent into the ACS terminal 408, exemplary operation of the simplifiedcircuit shown in FIG. 4C is the same as the operation of the circuitdescribed above with reference to FIG. 4B.

In yet another embodiment, the ACS terminal 408 may be coupled to acontrol circuit 412 as illustrated in the simplified circuit of FIG. 4D.The control circuit 412 may provide a selectable ACS bias voltageV_(ACS) that selectively controls the accumulated charge (i.e., theaccumulated charge 120 described above with reference to FIG. 1). Asshown in FIG. 4D, rather than having a local circuit provide the ACSbias voltage V_(ACS) (e.g., as derived from the gate voltage Vg), insome embodiments the ACS bias voltage V_(ACS) is produced by a separatesource that is independent of the ACC MOSFET device 300. In the case ofa switch (as described below in more detail with reference to FIG. 4E),the ACS bias voltage V_(ACS) should be driven from a source having ahigh output impedance. For example, such a high output impedance sourcecan be obtained using a large series resistor in order to ensure thatthe RF voltage is divided across the MOSFET and that the ACS biasvoltage V_(ACS) has Vds/2 “riding” on it, similarly to the gate voltage.This approach is described in more detail below with reference to FIG.4E.

It may be desirable to provide a negative ACS bias voltage V_(ACS) tothe ACS terminal 408 when the SOI NMOSFET 300 is biased into anaccumulated charge regime. In this exemplary embodiment, the controlcircuit 412 may prevent positive current flow into the ACS terminal 408by selectively maintaining an ACS bias voltage V_(ACS) that isconsistently negative with respect to both the source and drain biasvoltages. In particular, the control circuit 412 may be used to apply anACS bias voltage that is equal to or more negative than the lesser of Vsand Vd. By application of such an ACS bias voltage, the accumulatedcharge is thereby removed or otherwise controlled.

In the exemplary embodiment of the simplified circuit shown in FIG. 4D,the source and drain bias voltages applied to the terminals 404 and 406,respectively, may be zero. If the gate bias voltage (Vg) applied to thegate terminal 402 is sufficiently negative with respect to the sourceand drain bias voltages applied to the terminals 404 and 406, and withrespect to V_(th), (for example, if V_(th) is approximately zero, and ifVg is more negative than approximately −1 V) the ACC NMOSFET 300operates in the accumulated charge regime, and the accumulated charge(holes) may accumulate in the body of the ACC NMOSFET 300.Advantageously, the accumulated charge can be removed via the ACSterminal 408 by connecting the ACS terminal 408 to the control circuit412 as shown. In order to ensure that the accumulated charge is conveyedfrom the body of the ACC NMOSFET 300, the ACS bias voltage V_(ACS) thatis applied to the ACS terminal 408 should be equal to or more negativethan the gate voltage and more negative than the lesser of Vs and Vd.Because the accumulated charge 120 is conveyed to the bias voltageV_(ACS) applied to the ACS terminal 408 by the control circuit 412, theaccumulated charge does not remain trapped under the gate oxide due toattraction to the gate bias voltage Vg.

In other embodiments, Vs and Vd may comprise bias voltages that areother than zero. According to these examples, Vg must be sufficientlynegative to both Vs and Vd in order for Vg to be sufficiently negativeto V_(th), in order to bias the NMOSFET 300 in the off-state. Thisallows the accumulation of accumulated charge under the gate oxide. Forthis example, the ACS bias voltage V_(ACS) may be selected to be equalto or more negative than the lesser of Vs and Vd by connecting the ACSterminal 408 to the control circuit 412 to provide selected ACS biasvoltages, thereby conveying the accumulated charge from the ACC NMOSFET300.

In other embodiments, if the ACC NMOSFET 300 of FIG. 4D comprises adepletion mode device, V_(th) is, by definition, less than zero. For Vsand Vd both at zero volts, when a gate bias Vg sufficiently negative toV_(th) is applied (for example, Vg more negative than approximately −1 Vrelative to V_(th)), holes may accumulate under the gate oxide. For thisexample, the ACS bias voltage V_(ACS) that is applied to the ACSterminal 408 may also be selected to be equal to or more negative thanthe lesser of Vs and Vd by connecting the ACS terminal 408 to thecontrol circuit 412 and thereby provide the desired ACS bias voltagesV_(ACS) that are necessary to remove the accumulated charge from the ACCNMOSFET 300.

As described above, in one embodiment, instead of having the controlcircuit 412 provide a bias to the ACS terminal 408 as shown in FIG. 4D,the ACS terminal 408 can be driven by a separate bias source circuit, asshown, for example, in the embodiment of FIG. 4E. In one exemplarycircuit implementation, as exemplified in the circuit of FIG. 4E, in anRF switch circuit, the separate V_(ACS) source has a high outputimpedance element 403 which ensures that the RF voltage is dividedacross the ACC NMOSFET 300, and which further ensures that the voltageapplied to the ACS terminal 408 has Vds/2 applied thereon, similar tothe voltage Vgs that is applied to the gate terminal 402. In oneexemplary embodiment, an inverter 405 is configured in series with thehigh output impedance element 403 and supplied by GND and −V_(DD). Inone exemplary embodiment, −V_(DD) is readily derived from a convenientpositive voltage supply. It could, however, comprise an even morenegative voltage for improved linearity (i.e., it can be independent ofthe gate voltage).

In another embodiment, the circuit shown in FIG. 4C can be modified toinclude a clamping circuit configured in series with an ACS terminal408. Such an exemplary embodiment is shown in FIG. 4F. Under certainoperating conditions, current that flows out of the ACC NMOSFET 300,conveying the accumulated charge from the body of the ACC NMOSFET 300,via the ACS terminal 408 is sufficiently high such that it causesproblems in the biasing circuitry (i.e., under some conditions the ACScurrent is so high that the biasing circuitry cannot adequately sink thecurrent flowing out of the body of the ACC NMOSFET 300). As shown in thecircuit of FIG. 4F, one exemplary embodiment solves this problem byinterrupting the flow of ACS current out of the body of the ACC NMOSFET300, and thereby returning the ACC NMOSFET 300 to a floating bodycondition.

In one exemplary circuit, as shown in FIG. 4F, a depletion-mode FET 421is configured in series between the ACS terminal 408 and a diode 410. Inthis exemplary circuit, the depletion-mode FET 421 includes a gateterminal that is electrically connected to the FET's source terminal. Inthis configuration, the depletion-mode FET 421 functions to clip orlimit the current that flows from the ACS terminal 408 when the ACCMOSFET operates in the accumulated charge regime. More specifically, thedepletion-mode FET 421 enters saturation upon reaching a predefinedthreshold. The current leaving the body of the ACC MOSFET is therebylimited by the saturation current of the FET 421. In some embodiments,the predefined saturation threshold may optionally be adjusted to changethe point at which clamping occurs, such as by selecting a higherthreshold voltage, which results in a lower maximum current and earlierclamping.

In some embodiments, such as for example in an RF switch circuit, thegate terminal 402 and the ACS terminal 408 follow Vds at half the rate(Vds/2) of Vds. At high Vds excursions, Vgs may approach the thresholdvoltage Vth, resulting in increased Ids leakage current. In some cases,such a leakage current exits the ACS terminal 408 and can overwhelmassociated circuitry (e.g., a negative voltage generator). Hence, thecircuit shown in FIG. 4F solves or otherwise mitigates these problems.More specifically, by coupling the FET 421 in series between the ACSterminal 408 and the diode 410, the current that exits the ACS terminal408 is limited to the saturation current of the FET 421.

In yet another exemplary embodiment, the simplified circuit shown inFIG. 4C can be modified to include an AC shorting capacitor placed inparallel with the diode 410. The simplified circuit of FIG. 4G can beused to compensate for certain undesirable nonlinearities present in afull circuit application. In some embodiments, due to parasitics presentin the MOSFET layout, nonlinearity characteristics existing in the diode410 of FIG. 4C may introduce undesirable nonlinearities in a fullcircuit implementation. As the diode is in place to provide DC biasconditions and is not intended to have any AC signals across it, it maybe desirable in some embodiments to take steps to mitigate the effectsof any AC signal present across the diode 410.

As shown in the simplified circuit of FIG. 4G, the circuit of FIG. 4Chas been modified to include an AC shorting capacitor 423 wherein the ACshorting capacitor 423 is configured in parallel across the diode 410.The AC shorting capacitor 423 is placed in parallel with the diode 410to ensure that nonlinearities of the diode 410 are not excited by an ACsignal. In some exemplary circuits, such as in an RF switch, the ACshorting capacitor 423 does not impact the higher level full circuit, asthe gate terminal 402 and the ACS terminal 408 typically have the sameAC signal applied (i.e., AC equipotential).

In some circuit embodiments, body nodes of a multi-finger FETimplementation may be connected to one another (using, for example,metal or silicon), overlapping the source fingers. On another side ofthe FET implementation, gate nodes may be are connected to one another(using, for example, metal or silicon) overlapping the drain fingers. Asa result of this FET implementation, additional capacitance may resultbetween the source and body (S-B), and further additional capacitancemay result between the drain and gate (D-G). These additionalcapacitances may degrade the symmetry of the intrinsic device. Under ACexcitation, this results in the gate terminal following the drainterminal more closely, and the body terminal following the sourceterminal more closely, which effectively creates an AC signal across thediode 410, which can excite nonlinearities of the diode 410 as describedabove. Using the exemplary embodiment shown in FIG. 4G, parasiticnonlinear excitation due to the overlapping fingers is mitigated.

Improved C_(off) Performance Characteristics of ACC MOSFETs Made inAccordance with the Present Disclosed Method and Apparatus

FIG. 4H is a plot 460 of the off-state capacitance (C_(off)) versus anapplied drain-to-source voltage of an SOI MOSFET when an AC signal isapplied to the MOSFET (the plot 460 is relevant to an exemplary 1 mmwide MOSFET, though similar plots result using wider and narrowerdevices). In one embodiment, a gate voltage equals −2.5 Volts+Vd/2, andVs equals 0. A first plot 462 shows the off-state capacitance C_(off) ofa typical prior art NMOSFET operating within the accumulated chargeregime and thereby having an accumulated charge as described above withreference to FIG. 1. As shown in FIG. 4H, the off-state capacitanceC_(off) shown in plot 462 of the prior art FET is voltage-dependent(i.e., it is nonlinear) and peaks when Vd=0 Volts. A second plot 464illustrates the off-state capacitance C_(off) of an improved ACC SOIMOSFET made in accordance with the present teachings, wherein theaccumulated charge is conveyed from the ACC MOSFET, thereby reducing,controlling and/or eliminating the accumulated charge from the ACCMOSFET body. As shown in FIG. 4H, the off-state capacitance C_(off)shown in plot 464 of the ACC SOI MOSFET is not voltage-dependent (i.e.,it is linear).

As described above with reference to FIG. 2A, by controlling, reducingor eliminating the accumulated charge, the impedance 212 of the NMOSFETbody 312 (FIG. 3C, and shown as the MOSFET body 114 in the electricalmodel of FIG. 2A) is increased to a very large value. This increase inthe impedance 212 of the MOSFET body reduces the contribution to C_(off)caused by the impedance of the junctions 218 and 220 (FIG. 2A), therebyreducing the overall magnitude of C_(off) and the nonlinear effectsassociated with the impedances of the junctions 218 and 220. Plot 464illustrates how the present teachings advantageously reduce both thenonlinearity and overall magnitude of the off-state capacitance C_(off)of the MOSFET. The reduced nonlinearity and magnitude of the off-statecapacitance C_(off) improves the performance of circuits using MOSFETsoperating in an accumulated charge regime, such as RF switchingcircuits. Exemplary RF switching circuits implemented with the ACCMOSFETs described above with reference to FIGS. 4A-4G are now describedwith reference to FIGS. 5A-5D.

Exemplary Improved Performance RF Switch Implementations Using ACC SOIMOSFETs in Accordance with the Present Teachings

FIG. 5A shows a schematic diagram of a single pole, single throw (SPST)RF switch circuit 500 in accordance with prior art. The RF switchcircuit 500 is one example of a general class of well-known RF switchcircuits. Similar RF switch circuits are described in the followingmco-pending and commonly assigned U. S. Applications and Patent:Provisional Application No. 60/651,736, filed Feb. 9, 2005, entitled“UNPOWERED SWITCH AND BLEEDER CIRCUIT:” application Ser. No. 10/922,135,filed Aug. 18, 2004, pending, which is a continuation application ofapplication Ser. No. 10/267,531, filed Oct. 8, 2002, which issued Oct.12, 2004 as U.S. Pat. No. 6,804,502, entitled “SWITCH CIRCUIT AND METHODOF SWITCHING RADIO FREQUENCY SIGNALS”. Application Ser. No. 10/267,531,filed Oct. 8, 2002, which issued Oct. 12, 2004 as U.S. Pat. No.6,804,502 claims the benefit of U.S. Provisional Application No.60/328,353, filed Oct. 10, 2001. All of the above-cited applications andissued patent set forth above are hereby incorporated by referenceherein as if set forth in full for their teachings on RF switch circuitsincluding SOI MOSFET switch circuits.

Referring again to FIG. 5A, a switching SOI NMOSFET 506 is adapted toreceive an RF input signal “RFin” at an input terminal 502. Theswitching SOI MOSFET 506 is electrically coupled to selectively couplethe RFin input signal to an output terminal 504 (i.e., thereby convey anRF output signal Rfout at the output terminal 504). In the exemplaryembodiment, the switching SOI NMOSFET 506 is controlled by a firstcontrol signal C1 that is conveyed by a control line 512 through a gateresistor 510 (optionally included for suppression of parasitic RFcoupling). The control line 512 is electrically coupled to a controlcircuit 520, which generates the first control signal C1.

Referring again to FIG. 5A, a shunting SOI NMOSFET 508 is adapted toreceive the RF input signal RFin at its drain terminal, and toselectively shunt the input signal RFin to ground via an optional loadresistor 518. The shunting SOI NMOSFET 508 is controlled by a secondcontrol signal C1 x which is conveyed by a control line 516 through agate resistor 514 (optionally included for suppression of parasitic RFcoupling and for purposes of voltage division). The control line 516 iselectrically coupled to the control circuit 520, which generates thesecond control signal C1 x.

The terms “switching” and “shunting”, as pertains to the transistorsshown in FIG. 5A and also described below with reference to the RFswitch circuits of FIGS. 5B-5D, 6, 8, and 9, are used interchangeablyherein with the terms “switch” and “shunt”, respectively. For example,the switching transistor 506 (and all of its analogous switchingtransistors described below in FIGS. 5B-5D, 6, 8, and 9) is alsoreferred to herein as the “switch” transistor. Similarly, the shuntingtransistor 508 (and all of its analogous shunting transistors describedbelow in FIGS. 5B-5D, 6, 8, and 9) is also referred to herein as the“shunt” transistor. The terms “switch” and “switching” (and similarlythe terms “shunt” and “shunting”), when used to describe the RF switchcircuit transistors, are used interchangeably herein. Further, asdescribed below in more detail with reference to FIG. 6, those skilledin the RF switching design and fabrication arts shall recognize thatalthough the switch and shunt transistors are shown in FIGS. 5A-5D andFIG. 9 as comprising a single MOSFET, it shall be understood that theymay comprise transistor groupings comprising one or more MOSFETtransistors.

It will also be appreciated by those skilled in RF switch circuits thatall of the exemplary switch circuits may be used “bi-directionally,”wherein the previously described input ports function as output ports,and vice versa. That is, although an exemplary RF switch may bedescribed herein as having one or more input ports (or nodes) and one ormore output ports (or nodes), this description is for convenience only,and it will be understood that output ports may, in some applications,be used to input signals, and input ports may, in some applications, beused to output signals. The RF switch circuits described with referenceto FIGS. 2B, 4E, 5A-5D, 6, 8 and 9 are described herein as having“input” and “output” ports (or “nodes”) that input and output RFsignals, respectively. For example, as described below in more detailwith reference to FIG. 9, RF input node 905 and RF input node 907 aredescribed below as inputting RF signals RF1 and RF2 respectively. RFCcommon port 903 is described below as providing an RF common outputsignal. Those skilled in the RF switch circuit design arts shallrecognize that the RF switch is bidirectional, and that the previouslydescribed input ports function perfectly well as output ports, and viceversa. In the example of the RF switch of FIG. 9, the RFC common portcan be used to input an RF signal which is selectively output by the RFnodes 905 and 907.

Referring again to FIG. 5A, the first and second control signals, C1 andC1 x, respectively, are generated so that the switching SOI NMOSFET 506operates in an on-state when the shunting SOI NMOSFET 508 operates in anoff-state, and vice versa. These control signals provide the gate biasvoltages Vg to the gate terminals of the NMOSFETs 506 and 508. Wheneither of the NMOSFETs 506 or 508 is biased to select the transistoroff-state, the respective Vg must comprise a sufficiently large negativevoltage so that the respective NMOSFET does not enter, or approach, anon-state due to the time varying applied voltages of the RF input signalRFin. The maximum power of the RF input signal RFin is thereby limitedby the maximum magnitude of the gate bias voltage Vg (or, moregenerally, the gate-to-source operating voltage, Vgs) that the SOINMOSFETs 506 and 508 can reliably sustain. For RF switching circuitssuch as those exemplified herein, the magnitude ofVgs(max)=|Vg|+|Vds(max)/2|, where Vds=Vd−Vs, and Vds(max) comprises themaximum Vds due to the high-power input signal voltage levels associatedwith the RF input signal RFin.

Exemplary bias voltages for the switching and shunting SOI NMOSFETs 506and 508, respectively, may include the following: with V_(th)approximately zero volts, Vg, for the on-state, of +2.5 V, and Vg, forthe off-state, of −2.5 V. For these bias voltages, the SOI NMOSFETs mayeventually operate in an accumulated charge regime when placed intotheir off-states. In particular, and as described above with referenceto FIG. 2B, when the switching NMOSFET 506 is in the on-state, and theshunting NMOSFET 508 is biased in the off-state, the output signal RFoutmay become distorted by the nonlinear behavior of the off capacitanceC_(off) of the shunting NMOSFET 508 caused by the accumulated charge.Advantageously, the improved ACC MOSFETs made in accordance with thepresent teachings can be used to improve circuit performance, especiallyas it is adversely affected by the accumulated charge.

FIG. 5B is a schematic of an improved RF circuit 501 adapted for higherperformance using the present accumulated charge reduction and controltechniques. The switch circuit 501 differs from the prior art circuit500 (FIG. 5A) in that the shunting NMOSFET 508 is replaced by a shuntingACC NMOSFET 528 made in accordance with the present teachings. Theshunting ACC NMOSFET 528 is analogous to the ACC NMOSFET described abovewith reference to FIGS. 4A and 4B. Similarly, the gate, source, drainand ACC terminals of the shunting ACC NMOSFET 528 are analogous to therespective terminals of the ACC NMOSFET 300. With the exception of theimproved switch performance afforded by the improved shunting ACCNMOSFET 528, the operation of the RF switch circuit 501 is very similarto the operation of the RF switch circuit 500 described above withreference to FIG. 5A.

Exemplary bias voltages for the switching NMOSFET 526 and the shuntingACC NMOSFET 528 may include: with V_(th) approximately zero, Vg, for theon-state, of +2.5 V, and Vg, for the off-state, of −2.5 V. For thesebias voltages, the SOI NMOSFETs may operate in an accumulated chargeregime when placed into the off-state. However, when the switchingNMOSFET 526 is in the on-state and the shunting ACC NMOSFET 528 is inthe off-state, the output signal RFout at the output terminal 505 willnot be distorted by nonlinear behavior of the off-state capacitanceC_(off) of the improved shunting ACC NMOSFET 528 due to the accumulatedcharge. When the shunting ACC NMOSFET 528 operates in the accumulatedcharge regime, the accumulated charge is removed via the ACS terminal508′. More specifically, because the gate terminal 502′ of the shuntingACC NMOSFET 528 is connected to the ACS terminal 508′, the accumulatedcharge is removed or otherwise controlled as described above inreference to the simplified circuit of FIG. 4B. The control of theaccumulated charge improves performance of the switch 501 by improvingthe linearity of the off transistor, shunting ACC NMOSFET 528, andthereby reducing the harmonic and intermodulation distortion of the RFoutput signal Rfout generated at the output terminal 505.

FIG. 5C is a schematic of another embodiment of an improved RF switchcircuit 502 adapted for higher performance using the accumulated chargecontrol techniques of the present disclosure. The switch circuit 502differs from the prior art circuit 500 (FIG. 5A) in that the NMOSFET 508is replaced by an ACC NMOSFET 528 made in accordance with the presentteachings. The ACC NMOSFET 528 is analogous to the ACC NMOSFET 300described above with reference to FIGS. 4A and 4C. Similarly, the gate,source, drain and ACC terminals of the ACC NMOSFET 528 are analogous tothe respective terminals of the ACC NMOSFETs 300 described above withreference to FIGS. 4A and 4C. With the exception of the improved switchperformance afforded by the improved ACC NMOSFET 528, the operation ofthe switch circuit 502 is very similar to the operations of the switchcircuits 500 and 501 described above with reference to FIGS. 5A and 5B,respectively.

Exemplary bias voltages for the NMOSFET 526 and the ACC NMOSFET 528 mayinclude the following: with V_(th) approximately zero volts, Vg, for theon-state, of +2.5 V, and Vg, for the off-state, of −2.5 V. For thesebias voltages, the SOI NMOSFETs 526, 528 may operate in an accumulatedcharge regime when placed into an off-state. However, when the NMOSFET526 is in the on-state and the ACC NMOSFET 528 is in the off-state, theoutput signal RFout will not be distorted by nonlinear behavior of theoff-state capacitance C_(off) of the ACC NMOSFET 528 due to theaccumulated charge. Because the gate terminal 502′ of the ACC NMOSFET528 is connected to the ACS terminal 508′ via a diode 509, theaccumulated charge is entirely removed, reduced or otherwise controlled,as described above with reference to FIG. 4C. Similar to the improvedswitch 501 described above with reference to FIG. 5B, control of theaccumulated charge improves performance of the switch 502 by improvingthe linearity of the off transistor, 528, and thereby reducing theharmonic and intermodulation distortion of the RF output signal Rfoutoutput of the RF output terminal 505. Connection of the diode 509 asshown may be desired in some embodiments for suppression of positivecurrent flow into the ACC NMOSFET 528 when it is biased into anon-state, as described above with reference to FIG. 4C.

FIG. 5D is a schematic of another embodiment of an improved RF switchcircuit 503 adapted for higher performance using the present accumulatedcharge control techniques. The switch circuit 503 differs from the priorart circuit 500 (FIG. 5A) in that the NMOSFET 508 of FIG. 5A is replacedby an ACC NMOSFET 528 made in accordance with the present teachings. TheACC NMOSFET 528 is analogous to the ACC NMOSFET described above withreference to FIGS. 4A and 4D. With the exception of the improved switchperformance afforded by the improved ACC NMOSFET 528, the operation ofthe switch circuit 503 is very similar to the operations of the switchcircuits 500, 501 and 502 described above with reference to FIGS. 5A-5C,respectively.

Exemplary bias voltages for the NMOSFET 526 and the ACC NMOSFET 528 mayinclude the following: with V_(th) approximately zero volts, Vg, for theon-state, of +2.5 V, and Vg, for the off-state, of −2.5 V. For thesebias voltages, the SOI NMOSFETs 526, 528 may operate in an accumulatedcharge regime when placed into the off-state. However, when the NMOSFET526 is in the on-state and the ACC NMOSFET 528 is in the off-state, theoutput signal RFout produced at the output terminal 505 will not bedistorted by the nonlinear behavior of the off-state capacitance C_(off)of the ACC NMOSFET 528 due to the accumulated charge. When the NMOSFET528 operates in the accumulated charge regime, the accumulated charge isremoved via the ACS terminal 508′. More specifically, because the ACSterminal 508′ of the ACC NMOSFET 528 is electrically coupled to thecontrol circuit 520 via the control line 517 (i.e., controlled by thecontrol signal “C2” as shown), the accumulated charge can be eliminated,reduced or otherwise controlled by applying selected bias voltages tothe ACS terminal 508′ as described above with reference to FIG. 4D.Those skilled in the arts of electronic circuit design shall understandthat a wide variety of bias voltage signals can be applied to the ACSterminal for the purpose of reducing or otherwise controlling theaccumulated charge. The specific bias voltages may be adapted for use ina particular application. The control of the accumulated charge improvesperformance of the switch 503 by improving the linearity of theoff-state transistor, 528, and thereby reducing the harmonic andintermodulation distortion of the RF output signal Rfout generated atthe output terminal 505.

In the circuits described above with respect to FIGS. 5B-5D, theswitching SOI MOSFETs 526 are shown and described as implemented usingSOI MOSFETs of the prior art (i.e., they do not comprise ACC MOSFETs andtherefore do not have an ACS terminal). Those skilled in the electronicdevice design arts shall understand and appreciate that in otherembodiments of the disclosed method and apparatus, the prior artswitching SOI MOSFETs 526 may be replaced, as desired or required, byACC SOI MOSFETs made in accordance with the present disclosure. Forexample, in some embodiments of RF switches implemented using the ACCMOSFET of the present teachings, the RF switch comprises a single-poledouble-throw RF switch. In this embodiment, the switching SOI MOSFETs(e.g., analogous to the switching SOI MOSFETs 526 described above withreference to FIGS. 5B-5D) may comprise ACC SOI MOSFETs. Such animplementation prevents nonlinear behavior of the off-state switchingSOI MOSFETs (which is turned off when it is not selected as an input“pole”) from detrimentally affecting the output of the RF signal asswitched through the selected “pole”. Implementation of the RF switchesusing switching ACC MOSFETs reduces the magnitude, drift, and voltagedependency of the off capacitance C_(off) of the switching transistors.Consequently, and as described above in more detail, the switchperformance characteristics, such as its isolation, insertion loss anddrift characteristics, are also improved. This implementation isdescribed in more detail below with reference to the RF switch circuitshown in FIG. 9. Many other examples will be apparent to those skilledin the arts of electronic circuits.

For example, as set forth above, although the exemplary RF switches havebeen described as being implemented using ACC SOI NMOSFET devices, theycan also be implemented using ACC SOI PMOSFET devices. Further, althoughsingle-pole single-throw, and single-pole double-throw RF switches havebeen described above as examples of RF switches implemented inaccordance with the present teachings, the present applicationencompasses any variation of single-pole multi-throw, multi-polesingle-throw, and multi-pole multi-throw RF switch configurations. Thoseskilled in the RF switch design and fabrication arts shall recognize andappreciate that the present teachings can be used in implementing anyconvenient RF switch configuration design.

Exemplary RF Switch Implementation Using Stacked Transistors

In the exemplary embodiments of RF switch circuits described above, theswitch circuits are implemented using a single SOI NMOSFET (e.g., thesingle SOI NMOSFET 506 of FIG. 5A, and the single SOI NMOSFET 526 ofFIGS. 5B-5D) that selectively couples or blocks (i.e., electricallyopens the circuit connection) the RF input signal to the RF output.Similarly, in the exemplary embodiments described above with referenceto FIGS. 5A-5D, a single SOI NMOSFET (e.g., the single SOI NMOSFET 508of FIG. 5A, and ACC SOI NMOSFET 528 of FIGS. 5B-5D) is used to shunt(FET in the on-state) or block (FET in the off-state) the RF inputsignal to ground. Commonly assigned U.S. Pat. No. 6,804,502, entitled“SWITCH CIRCUIT AND METHOD OF SWITCHING RADIO FREQUENCY SIGNALS”, issuedOct. 12, 2004, describes RF switch circuits using SOI NMOSFETsimplemented with stacked transistor groupings that selectively coupleand block RF signals.

One example of how stacked NMOSFETs may be implemented in accordancewith the teachings of the present disclosure is illustrated in FIG. 6.An RF switch circuit 600 is analogous to the RF switch circuit 503 ofFIG. 5D, wherein the single SOI NMOSFET 526 is replaced by a stack ofSOI NMOSFETs 602, 604 and 606. Similarly, the single ACC SOI NMOSFET 528is replaced by a stack of ACC SOI NMOSFETs 620, 622 and 624. The controlsignal C2 is provided to the ACS terminals of the ACC SOI NMOSFETs 620,622 and 624 via optional resistors 626, 628, and 630, respectively. Theresistors 626, 628, and 630 may optionally be included in order tosuppress parasitic RF signals between the stacked ACC SOI NMOSFETs 620,622, and 624, respectively. The RF switch circuit 600 operatesanalogously to the operation of the RF switch circuit 503 describedabove with reference to FIG. 5D.

Three stacked ACC SOI NMOSFETs are shown in each ACC NMOSFET stack inthe exemplary stacked RF switch circuit 600 of FIG. 6. A plurality ofthree ACC NMOSFETs is shown for illustrative purposes only, however,those skilled in the integrated circuit design arts will understand thatan arbitrary plurality may be employed according to particular circuitrequirements such as power handling performance, switching speed, etc. Asmaller or larger plurality of stacked ACC NMOSFETs may be included in astack to achieve a desired operating performance.

Other stacked RF switch circuits, adapted for accumulated chargecontrol, analogous to the circuits described above with reference toFIGS. 5B-5D, may also be employed. Implementations of such circuitsshall be obvious from the teachings above to those skilled in theelectronic device design arts, and therefore are not described furtherherein. Moreover, is shall be obvious to those skilled in the electronicdevice design arts that, although a symmetrically stacked (i.e., havingan equal number of shunting and switching transistors) RF switch isshown in the stacked RF switch of FIG. 6, the present inventive ACCmethod and apparatus is not so limited. The present teachings can beapplied in implementing both symmetrically and asymmetrically stacked(having an unequal number of shunting and switching transistors) RFswitches. The designer will readily understand how to use the ACCMOSFETs of the present disclosure in implementing asymmetrical, as wellas symmetrical, RF switch circuits.

Exemplary Method of Operation

FIG. 7 illustrates an exemplary method 700 of improving the linearity ofan SOI MOSFET having an accumulated charge sink (ACS) in accordance withthe present disclosure. The method 700 begins at a STEP 702, whereat anACC SOI MOSFET having an ACS terminal is configured to operate in acircuit. The ACS terminal may be operatively coupled to the gate of theSOI MOSFET (as described above with reference to FIGS. 4B, 4C, 5B and5C), or to a control circuit (as described above with reference to FIGS.4D and 5D). In other embodiments, the ACS terminal may be operativelycoupled to any convenient accumulated charge sinking mechanism, circuit,or device as is convenient to the circuit or system designer. The methodthen proceeds to a step 704.

At the STEP 704, the ACC SOI MOSFET is controlled, at least part of thetime, so that it operates in an accumulated charge regime. In mostembodiments, as described above, the ACC MOSFET is operated in theaccumulated charge regime by applying bias voltages that place the FETin an off-state condition. In one exemplary embodiment, the ACC SOIMOSFET comprises an ACC SOI NMOSFET that is configured as part of ashunting circuit of an RF switch. According to this exemplaryembodiment, the SOI NMOSFET may be operated in an accumulated chargeregime after the shunting circuit is placed into an off-state byapplying a negative bias voltage to the gate terminal of the ACCNMOSFET.

The method then proceeds to a STEP 706, whereat the accumulated chargethat has accumulated in the channel region of the ACC MOSFET is removedor otherwise controlled via the ACS terminal. In this embodiment, theaccumulated charge is conveyed to another circuit terminal and isthereby reduced or otherwise controlled. One such exemplary circuitterminal that can be used to convey the accumulated charge from theMOSFET body comprises a gate terminal of the ACC MOSFET (see, e.g., thedescription above with reference to FIGS. 4B, 4C, 5B and 5C). Anotherexemplary circuit terminal that can be used to remove or otherwisecontrol the accumulated charge comprises the terminal of a controlcircuit (see, e.g., FIGS. 4D and 5D). As described in more detail above,removing or otherwise controlling the accumulated charge in the ACCMOSFET body improves the linearity of the off-state ACC MOSFET, whichreduces the harmonic distortion and IMD of signals affected by the ACCMOSFET, and which, in turn, improves circuit and system performance. InRF switch circuits, improvements (in both linearity and magnitude) aremade to the off capacitance of shunting ACC MOSFET devices, which, inturn, improves the performance of the RF switch circuits. In addition toother switch performance characteristics, the harmonic andintermodulation distortions of the RF switch are reduced using the ACCmethod and apparatus of the present teachings.

FIGS. 8 and 9 show schematics of additional exemplary embodiments of RFswitching circuits made in accordance with the disclosed method andapparatus for use in improving linearity of MOSFETs having an ACS. Asdescribed in more detail below with reference to FIGS. 8 and 9, in someexemplary embodiments of RF switch circuits made in accordance with thepresent disclosure, it may be desirable to include drain-to-sourceresistors, Rds, and thereby improve some switch performancecharacteristics when the switch is used in a particular application.These exemplary RF switch circuits are now described in more detail.

Exemplary RF Switch Implementations Using Stacked Transistors HavingSource to Drain Resistors

FIG. 8 shows one exemplary embodiment of an RF switch circuit 800 madein accordance with the present disclosure. As shown in FIG. 8, someembodiments of RF switches made in accordance with the presentdisclosure may include drain-to-source (R_(ds)) resistors electricallyconnected to the respective sources and drains of the ACC MOSFETs. Forexample, the exemplary switch 800 of FIG. 8 includes drain-to-sourceR_(ds) resistors 802, 804, and 806 electrically connected to therespective sources and drains of the shunting ACC SOI NMOSFETs 620, 622,and 624, respectively. Motivation for use of the drain-to-source R_(ds)resistors is now described.

As shall be appreciated by skilled persons from the present teachings,removal of the accumulated charge via the ACS terminal causes current toflow from the body of the ACC SOI MOSFET. For example, when a holecurrent flows from the body of an ACC SOI MOSFET via the ACS, an equalelectron current flows to the FET source and/or drain. For some circuits(e.g., the RF switch circuit of FIG. 8), the sources and/or drains ofthe ACC SOI NMOSFETs are connected to other SOI NMOSFETs. Becauseoff-state SOI NMOSFETs have a very high impedance (e.g., in the range of1 Gohm for a 1 mm wide SOI NMOSFET), even a very small drain-to-sourcecurrent (e.g., in the range of 1 nA) can result in an unacceptably largedrain-to-source voltage Vds across the ACC SOI NMOSFET in satisfactionof Kirchhoff's well known current and voltage laws. In some embodiments,such as that shown in the RF switch circuits of FIGS. 8 and 9, suchresultant very large drain-to-source voltages Vds undesirably impactsreliability and linearity of the ACC SOI NMOSFET. The drain-to-sourceresistors R_(ds) provide a path between the ACC FET drain and sourcewhereby currents associated with controlling the accumulated charge maybe conducted away from the sources and drains of ACC SOI NMOSFETs whenimplemented in series with high impedance elements such as other ACC SOINMOSFETs.

Exemplary operating voltages for the NMOSFETs 602-606 of FIG. 8, and theACC NMOSFETs 620-624, may include the following: V_(th) approximatelyzero volts, Vg, for the on-state, of +2.5 V, and Vg, for the off-state,of −2.5 V. In an exemplary embodiment, the ACC SOI NMOSFET 622 of FIG. 8may have a width of 1 mm, and an electron-hole pair generation rate foraccumulated charge producing a current of 10 pA/μm for operation in theaccumulated charge regime. For the electron current supplied equally bythe source and drain, and an impedance of the ACC SOI NMOSFETs 620 and622 on the order of 1 Gohm, then an unacceptable bias of −5 V wouldresult on the source and drain of the ACC SOI NMOSFET 622 without thepresence of R_(ds) resistors 802 and 806. This bias voltage would alsobe applied to the interior nodes of the ACC SOI NMOSFETs 620 and 624.

Even currents smaller than the exemplary currents may produce adverseaffects on the operation of the RF switching circuit 800 by reducing Vgsand/or Vgd of the ACC SOI MOSFETs 620-624 in the off-state, therebyreducing the power handling capability and reliability of the circuit byincreasing leakage (e.g., when either Vgs or Vgd approaches V_(th)), byincreasing hot-carrier damage caused by excess leakage, etc. Linearityof the MOSFETs is also degraded by reducing Vgs and/or Vgd when eithervalue approaches V_(th).

Exemplary values for the R_(ds) resistors 802 to 806 may be selected insome embodiments by selecting a value approximately equal to theresistance of the gate resistors 632-636 divided by the number of ACCSOI NMOSFETs in the stack (in the exemplary embodiment, there are threeACC FETs in the stack). More generally, the value of the R_(ds)resistors may be equal to the gate resistor value divided by the numberof ACC SOI NMOSFETs in the stack. In one example, a stack of eight ACCSOI NMOSFETs may have gate resistors of 80 kohm and R_(ds) resistors of10 kohm.

In some embodiments, the R_(ds) resistors may be selected so that theydo not adversely affect switch performance characteristics, such as, forexample, the insertion loss of the switch 800 due to the off-state ACCSOI NMOSFETs. For example, for a net shunt resistance greater than 10kohm, the insertion loss is increased by less than 0.02 dB.

In other embodiments, the R_(ds) resistors may be implemented incircuits comprising a single ACC SOI MOSFET (as contrasted with thestacked shunting configuration exemplified in FIG. 8 by the shunting ACCFETs 620, 622 and 624). For example, such circuits may be desirable ifthere are other high-impedance elements configured in series with an ACCSOI MOSFET that may cause a significant bias voltage to be applied tothe source or drain as a result of the current flow created whenremoving or otherwise controlling accumulated charge. One exemplaryembodiment of such a circuit is shown in FIG. 9.

FIG. 9 shows an exemplary single-pole double-throw (SPDT) RF switchcircuit 900 made in accordance with the present teachings. As shown inFIG. 9, a DC blocking capacitor 904 is connected to a first RF inputnode 905 that receives a first RF input signal RF1. Similarly, a DCblocking capacitor 906 is connected to a second RF input node 907 thatreceives a second RF input signal RF2. Further, a DC blocking capacitor902 is electrically connected to an RF common output node 903 thatprovides an RF common output signal (RFC) selectively conveyed to thenode RFC 903 by the switch circuit 900 from either the first RF inputnode 905 or the second RF input node 907 (i.e., RFC either outputs RF1or RF2, depending upon the operation of the switch as controlled by thecontrol signals C1 and C1 x described below in more detail).

A first control signal C1 is provided to control the operating states ofthe ACC SOI NMOSFETs 526 and 528′ (i.e., C1 selectively operates theFETs in the on-state or the off-state). Similarly, a second controlsignal C1 x is provided to control the operating states of the ACC SOINMOSFETs 528 and 526′. As is well known, and as described for example inthe above incorporated commonly assigned U.S. Pat. No. 6,804,502, thecontrol signals C1 and C1 x are generated so that the ACC SOI NMOSFETs526 and 528′ are in an on-state when the ACC SOI NMOSFETs 528 and 526′are in an off-state, and vice versa. This configuration allows the RFswitch circuit 900 to selectively convey either the signal RF1 or RF2 tothe RF common output node 903.

A first ACS control signal C2 is configured to control the operation ofthe ACS terminals of the SOI NMOSFETs 526 and 528′. A second ACS controlsignal C2 x is configured to control the ACS terminals of the ACC SOINMOSFETs 528 and 526′. The first and second ACS control signals, C2 andC2 x, respectively, are selected so that the ACSs of the associated andrespective NMOSFETs are appropriately biased in order to eliminate,reduce, or otherwise control their accumulated charge when the ACC SOINMOSFETs operate in an accumulated charge regime.

As shown in the RF switch circuit 900 of FIG. 9, in some embodiments, anR_(ds) resistor 908 is electrically connected between the source anddrain of the switching ACC NMOSFET 526. Similarly, in some embodiments,an R_(ds) resistor 910 is electrically connected between the source anddrain of the switching ACC NMOSFET 526′. According to this example, thecircuit 900 is operated so that either the shunting ACC NMOSFET 528 orthe shunting ACC NMOSFET 528′ operate in an on-state at any time (i.e.,at least one of the input signals RF1 at the node 905 or RF2 at the node907 is always conveyed to the RFC node 903), thereby providing alow-impedance path to ground for the node 905 or 907, respectively.Consequently, either the R_(ds) resistor 908 or the R_(ds) resistor 910provides a low-impedance path to ground from the RF common node 903,thereby preventing voltage bias problems caused as a result of ACCcurrent flow into the nodes 903, 905 and 907 that might otherwise becaused when using the DC blocking capacitors 902, 904 and 906.

Additional Exemplary Benefits Afforded by the ACC MOSFETs of the PresentDisclosure

As described above, presence of the accumulated charge in the bodies ofthe SOI MOSFETs can adversely affect the drain-to-source breakdownvoltage (BVDSS) performance characteristics of the floating bodyMOSFETs. This also has the undesirable effect of worsening the linearityof off-state MOSFETs when used in certain circuits such as RF switchingcircuits. For example, consider the shunting SOI NMOSFET 528 shown inFIG. 9. Further consider the case wherein the shunting NMOSFET 528 isimplemented with a prior art SOI NMOSFET, rather than with the ACCNMOSFET made in accordance with the present teachings. Assume that theRF transmission line uses a 50-ohm system. With small signal inputs, andwhen the NMOSFET 528 operates in an off-state, the prior art off-stateshunting NMOSFET 528 may introduce harmonic distortion and/orintermodulation distortion in the presence of multiple RF signals Thiswill also introduce a noticeable loss of signal power.

When sufficiently large signals are input that cause the NMOSFET 528 toenter a BVDSS regime, some of the RF current is clipped, or redirectedthrough the NMOSFET 528 to ground, resulting in a loss of signal power.This current “clipping” causes compression behavior that can be shown,for instance, in a RF switch “Pout vs. Pin” plot. This is frequentlycharacterized by P1 dB, wherein the insertion loss is increased by 1.0dB over the small-signal insertion loss. This is an obvious indicationof nonlinearity of the switch. In accordance with the present disclosedmethod and apparatus, removing, reducing or otherwise controlling theaccumulated charge increases the BVDSS point. Increases to the BVDSSpoint of the NMOSFET 528 commensurately increases the large-signal powerhandling of the switch. As an example, for a switch, doubling the BVDSSvoltage of the ACC NMOSFET increases the P1 dB point by 6 dB. This is asignificant accomplishment as compared with the prior art RF switchdesigns.

In addition, as described above in more detail, presence of theaccumulated charge in SOI MOSFET body adversely impacts the magnitude ofC_(off) and also takes time to form when the FET is switched from anon-state to an off-state. In terms of switch performance, thenonlinearity of C_(off) adversely impacts the overall switch linearityperformance (as described above), and the magnitude of C_(off) adverselyaffects the small-signal performance parameters such as insertion loss,insertion phase (or delay), and isolation. By reducing the magnitude ofC_(off) using the present disclosed method and apparatus, the switch(implemented with ACC MOSFETs) has reduced insertion loss due to loweredparasitic capacitance, reduced insertion phase (or delay), again due tolowered parasitic capacitance, and increased isolation due to lesscapacitive feedthrough.

The ACC MOSFET also improves the drift characteristic of SOI MOSFETs aspertains to the drift of the small-signal parameters over a period oftime. As the SOI MOSFET takes some time to accumulate the accumulatedcharge when the switch is off, the C_(off) capacitance is initiallyfairly small. However, over a period of time while operated in theaccumulated charge regime, the off-state capacitance C_(off) increasestoward a final value. The time it takes for the NMOSFET to reach a fullaccumulated charge state depends on the electron-hole pair (EHP)generation mechanism. Typically, this time period is on the order ofapproximately hundreds of milliseconds for thermal EHP generation atroom temperature, for example. During this charge-up time period, theinsertion loss and insertion phase increase. Also, during this timeperiod, the isolation decreases. As is well known, these are undesirablephenomena in standard SOI MOSFET devices. These problems are alleviatedor otherwise mitigated using the ACC NMOSFETs and related circuitsdescribed above.

In addition to the above-described benefits afforded by the disclosedACC MOSFET method and apparatus, the disclosed techniques also allow theimplementation of SOI MOSFETs having improved temperature performance,improved sensitivity to Vdd variations, and improved sensitivity toprocess variations. Other improvements to the prior art SOI MOSFETsafforded by the present disclosed method and apparatus will beunderstood and appreciated by those skilled in the electronic devicedesign and manufacturing arts.

Exemplary Fabrication Methods

In one embodiment of the present disclosure, the exemplary RF switchesdescribed above may be implemented using a fully insulating substratesemiconductor-on-insulator (SOI) technology. Also, as noted above, inaddition to the commonly used silicon-based systems, some embodiments ofthe present disclosure may be implemented using silicon-germanium(SiGe), wherein the SiGe is used equivalently in place of silicon.

In some exemplary embodiments, the MOSFET transistors of the presentdisclosure may be implemented using “Ultra-Thin-Silicon (UTSi)” (alsoreferred to herein as “ultrathin silicon-on-sapphire”) technology. Inaccordance with UTSi manufacturing methods, the transistors used toimplement the inventive methods disclosed herein are formed in anextremely thin layer of silicon in an insulating sapphire wafer. Thefully insulating sapphire substrate enhances the performancecharacteristics of the inventive RF circuits by reducing the deleterioussubstrate coupling effects associated with non-insulating and partiallyinsulating substrates. For example, insertion loss improvements may berealized by lowering the transistor on-state resistances and by reducingparasitic substrate conductance and capacitance. In addition, switchisolation is improved using the fully insulating substrates provided byUTSi technology. Owing to the fully insulating nature ofsilicon-on-sapphire technology, the parasitic capacitance between thenodes of the RF switches is greatly reduced as compared with bulk CMOSand other traditional integrated circuit manufacturing technologies.

Examples of and methods for making silicon-on-sapphire devices that canbe implemented in the MOSFETs and circuits described herein, aredescribed in U.S. Pat. No. 5,416,043 (“Minimum charge FET fabricated onan ultrathin silicon on sapphire wafer”); U.S. Pat. No. 5,492,857(“High-frequency wireless communication system on a single ultrathinsilicon on sapphire chip”); U.S. Pat. No. 5,572,040 (“High-frequencywireless communication system on a single ultrathin silicon on sapphirechip”); U.S. Pat. No. 5,596,205 (“High-frequency wireless communicationsystem on a single ultrathin silicon on sapphire chip”); U.S. Pat. No.5,600,169 (“Minimum charge FET fabricated on an ultrathin silicon onsapphire wafer”); U.S. Pat. No. 5,663,570 (“High-frequency wirelesscommunication system on a single ultrathin silicon on sapphire chip”);U.S. Pat. No. 5,861,336 (“High-frequency wireless communication systemon a single ultrathin silicon on sapphire chip”); U.S. Pat. No.5,863,823 (“Self-aligned edge control in silicon on insulator”); U.S.Pat. No. 5,883,396 (“High-frequency wireless communication system on asingle ultrathin silicon on sapphire chip”); U.S. Pat. No. 5,895,957(“Minimum charge FET fabricated on an ultrathin silicon on sapphirewafer”); U.S. Pat. No. 5,920,233 (“Phase locked loop including asampling circuit for reducing spurious side bands”); U.S. Pat. No.5,930,638 (“Method of making a low parasitic resistor on ultrathinsilicon on insulator”); U.S. Pat. No. 5,973,363 (“CMOS circuitry withshortened P-channel length on ultrathin silicon on insulator”); U.S.Pat. No. 5,973,382 (“Capacitor on ultrathin semiconductor oninsulator”); and U.S. Pat. No. 6,057,555 (“High-frequency wirelesscommunication system on a single ultrathin silicon on sapphire chip”).All of these referenced patents are incorporated herein in theirentirety for their teachings on ultrathin silicon-on-sapphire integratedcircuit design and fabrication.

Similarly to other bulk and SOI CMOS processes, an SOS enhancement modeNMOSFET, suitable for some embodiments of the present disclosure, may,in some embodiments, be fabricated with a p-type implant into thechannel region with n-type source and drain regions, and may have athreshold voltage of approximately +500 mV. The threshold voltage isdirectly related to the p-type doping level, with higher dopingresulting in higher thresholds. Similarly, the SOS enhancement modePMOSFET may, in some exemplary embodiments, be implemented with ann-type channel region and p-type source and drain regions. Again, thedoping level defines the threshold voltage with higher doping resultingin a more negative threshold.

In some exemplary embodiments, an SOS depletion-mode NMOSFET, suitablefor some embodiments of the present disclosure, may be fabricated byapplying the p-type channel-implant mask to the n-type transistor,resulting in a structure that has n-type channel, source, and drainregions and a negative threshold voltage of approximately −500 mV.Similarly, in some exemplary embodiments, a suitable depletion-modePMOSFET may be implemented by applying the n-type channel-implant maskto the p-type transistor, resulting in a structure that has p-typechannel, source, and drain regions and a positive threshold voltage ofapproximately +500 mV.

As noted in the background section above, the present ACC MOSFETapparatus can also be implemented using any convenientsemiconductor-on-insulator technology, included, but not limited to,silicon-on-insulator, silicon-on-sapphire, and silicon-on-bonded wafertechnology. One such silicon-on-bonded wafer technique uses “directsilicon bonded” (DSB) substrates. Direct silicon bond (DSB) substratesare fabricated by bonding and electrically attaching a film ofsingle-crystal silicon of differing crystal orientation onto a basesubstrate. Such implementations are available from the Silicon GenesisCorporation headquartered in San Jose, Calif. As described at theSilicon Genesis Corporation website (publicly available at“www.sigen.com”), silicon-on-bonded wafer techniques include theso-called NanoCleave™ bonding process which can be performed at roomtemperature. Using this process, SOI wafers can be formed with materialshaving substantially different thermal expansion coefficients, such asin the manufacture of Germanium-on-Insulator wafers (GeOI). Exemplarypatents describing silicon-on-bonded wafer implementations are asfollows: U.S. Pat. No. 7,056,808, issued Jun. 6, 2006 to Henley, et al.;U.S. Pat. No. 6,969,668, issued Nov. 29, 2005 to Kang, et al.; U.S. Pat.No. 6,908,832, issued Jun. 21, 2005 to Farrens et al.; U.S. Pat. No.6,632,724, issued Oct. 14, 2003 to Henley, et al. and U.S. Pat. No.6,790,747, issued Sep. 14, 2004 to Henley, et al. All of the above-citedpatents are incorporated by reference herein for their teachings ontechniques and methods of fabricating silicon devices on bonded wafers.

A reference relating to the fabrication of enhancement-mode anddepletion-mode transistors in SOS is “CMOS/SOS/LSI Switching RegulatorControl Device,” Orndorff, R. and Butcher, D., Solid-State CircuitsConference, Digest of Technical Papers, 1978 IEEE International, VolumeXXI, pp. 234-235, February 1978. The “Orndorff” reference is herebyincorporated in its entirety herein for its techniques on thefabrication of enhancement-mode and depletion-mode SOS transistors.

Exemplary Results—Appendix A

Exemplary results that can be obtained using the disclosed method andapparatus for use in improving the linearity of MOSFETs are described inthe attached Appendix A, entitled “Exemplary Performance Results of anSP6T Switch Implemented with ACC MOSFETs”. The contents of Appendix Aare hereby incorporated by reference herein in its entirety. The resultsshown in detail in Appendix A are now briefly described. As noted in theattached Appendix A, the measured results are provided for a singlepole, six throw (SP6T) RF switch. Those skilled in the art of RF switchcircuit design shall understand that the results can be extended to anypractical RF switch configuration, and therefore are not limited to theexemplary SP6T switch for which results are shown.

Slides 2-7 of Appendix A show harmonic performance versus Input Powerfor prior art devices and for ACC MOSFET devices made in accordance withthe present disclosed method and apparatus. Switch circuits implementedwith the ACC MOSFET of the disclosed method and apparatus have a thirdharmonic response that rises at a 3:1 slope (cube of the input) versusinput power on the log scale. Those skilled in the electronic devicedesign arts shall appreciate that no input-power dependent dynamicbiasing occurs with the improved RF switch designs made in accordancewith the present disclosure. In contrast, prior art floating body FETharmonics disadvantageously do not follow a 3:1 slope. This isdisadvantageous for small-signal third-order distortions such as IM3.

As shown in the attached Appendix A, at a GSM maximum input power of +35dBm, the 3fo is improved by 14 dB. This is shown in detail in slidenumber 3 of the attached Appendix A. Improvements in third orderharmonic distortion is also applicable to all odd order responses, suchas, for example, 5^(th) order responses, 7^(th) order responses, etc.

Similar to 3fo, the second order response of the improved ACCMOSFET-implemented RF switch follows a 2:1 slope (square of the input)whereas the prior art RF switch does not. This results in improved 2foand IM2 performance at low input power, and roughly the same performanceat +35 dBm.

Slide numbers 6 and 7 of the attached Appendix A treat the performanceunder non 50-ohm loads. In this case, the load represents a 5:1 mismatchwherein the load impedance can be of any convenient value that resultsin a reflection coefficient magnitude of 2/3. In the case of SOIMOSFETs, reflection coefficients that result in higher voltages causethe most severe problems. At 5:1 VSWR, the voltage can be 1.667× higher.Those skilled in the art may view this similarly by sweeping the inputpower up to higher voltages which equate to the mismatch conditions.

The slides provided in Appendix A illustrate that the improved RFswitch, implemented with ACC MOSFETs made in accordance with the presentdisclosed method and apparatus, has improved large voltage handlingcapabilities as compared to the prior art RF switch implementations. Asshown in the slides, the harmonics are approximately 20 dB at the worstmismatch phase angle. Transient harmonics are also shown. Those skilledin the art shall observe that the standard SP6T switch 3fo overshoots byseveral dB before reaching a final value. The improved SP6T switch madein accordance with the present disclosed method and apparatus does notexhibit such a time-dependency.

Slide number 8 of the Appendix A shows insertion loss performanceresults achieved using the improved SP6T RF switch of the presentteachings. It can be observed that the improved SP6T switch has slightlyimproved insertion loss (IL) performance characteristics. Slide number 9of Appendix A shows that isolation is also slightly improved using thepresent improved SP6T RF switch.

Slide number 10 of the Appendix A shows IM3 performance which is ametric of the slightly nonlinear behavior of the RF switch. The IM3performance is shown versus phase again due to a load mismatch in thesystem under test. As can be observed by reviewing Slide number 10 ofthe Appendix A, the performance of the improved SP6T RF switch isimproved by 27 dB.

Finally, Slide number 11 of the Appendix A is a summary table which alsoincludes IM2 data. Slide number 11 shows almost 20 dB improvement for alow frequency blocker and 11 dB for a high frequency blocker. In oneexemplary application wherein the SP6T may be used, all IM products mustfall below −105 dBm. The improved SP6T switch is the only RF switchmanufactured at the time of filing the present application meeting thisrequirement.

A number of embodiments of the present inventive concept have beendescribed. Nevertheless, it will be understood that variousmodifications may be made without departing from the scope of theinventive teachings. For example, it should be understood that thefunctions described as being part of one module may in general beperformed equivalently in another module. Also, as described above, allof the RF switch circuits can be used in bi-directionally, with outputports used to input signals, and vice versa. Furthermore, the presentinventive teachings can be used in the implementation of any circuitthat will benefit from the removal of accumulated charge from MOSFETbodies. The present teachings will also find utility in circuits whereinoff-state transistors must withstand relatively high voltages. Otherexemplary circuits include DC-to-DC converter circuits, poweramplifiers, and similar electronic circuits.

Accordingly, it is to be understood that the concepts described hereinare not to be limited by the specific illustrated embodiments, but onlyby the scope of the appended claims.

What is claimed is:
 1. An RF module comprising: at least one integratedcircuit chip; the at least one integrated circuit chip included in theRF module and further including at least one field effect transistor,the at least one field effect transistor including a gate, a drain, asource, and a body; wherein, during at least a portion of an off state,the body of the at least one field effect transistor is to beelectrically biased to have a voltage level substantially more negativethan the lowest voltage level of the following: ground, the DC voltagelevel of the source of the at least one field effect transistor, and theDC voltage level of the drain of the at least one field effecttransistor.
 2. The RF module of claim 1, wherein the at least one fieldeffect transistor comprises a N-type metal oxide semiconductor (NMOS)field effect transistor.
 3. The RF module of claim 1, wherein the atleast one field effect transistor is implemented in a silicon oninsulator technology.
 4. The RF module of claim 1, wherein the voltagelevel substantially more negative than the lowest voltage level is morethan one volt more negative than the lowest voltage level.
 5. The RFmodule of claim 1, wherein the body of the at least one field effecttransistor is electrically coupled to an accumulated charge sink.
 6. TheRF module of claim 1, wherein the at least one field effect transistorduring the at least a portion of the off state is to be electricallybiased so as to improve the linearity of the at least one field effecttransistor relative to the at least one field effect transistor notelectrically biased to have a voltage level substantially more negativethan the lowest voltage level of the following: ground, the DC voltagelevel of the source of the at least one field effect transistor, and theDC voltage level of the drain of the at least one field effecttransistor.
 7. The RF module of claim 1, wherein the at least one fieldeffect transistor during the at least a portion of the off state is tobe electrically biased so as to reduce non-linear harmonic and/orintermodulation distortion of RF signals to be propagated by the module,reduction via the at least one field effect transistor being relative tothe at least one field effect transistor not electrically biased to havea voltage level substantially more negative than the lowest voltagelevel of the following: ground, the DC voltage level of the source ofthe at least one field effect transistor, and the DC voltage level ofthe drain of the at least one field effect transistor.
 8. The RF moduleof claim 7, wherein the power of a third harmonic of the RF signals tobe propagated via the RF module is to be lower than −30 dBm at anoperating power of +35 dBm power.
 9. The RF module of claim 1, whereinthe at least one field effect transistor is included in a stack of fieldeffect transistors included in an RF switch, the RF switch beingincluded in the at least one integrated circuit chip.
 10. The RF moduleof claim 1, wherein the body of the at least one field effect transistoris electrically coupled to at least two accumulated charge sinks.
 11. Acommunication device comprising: at least one integrated circuit chip;the at least one integrated circuit chip is included in thecommunication device and further including an RF switch comprising atleast one field effect transistor, the at least one field effecttransistor includes a gate, a drain, a source, and a body; whereinduring at least a portion of an off state, the body of the at least onefield effect transistor is to be electrically biased to have a voltagelevel substantially more negative than the lowest voltage level of thefollowing: ground, the DC voltage level of the source of the at leastone field effect transistor, and the DC voltage level of the drain ofthe at least one field effect transistor.
 12. The RF module of claim 11,wherein the at least one field effect transistor comprises a N-typemetal oxide semiconductor (NMOS) field effect transistor.
 13. Thecommunication device of claim 11, wherein the at least one field effecttransistor is implemented in a silicon on insulator technology.
 14. Thecommunication device of claim 11, wherein the voltage levelsubstantially more negative than the lowest voltage level is more thanone volt more negative than the lowest voltage level.
 15. Thecommunication device of claim 11, wherein, the body of the at least onefield effect transistor is electrically coupled to an accumulated chargesink.
 16. The communication device of claim 11, wherein the at least onefield effect transistor during the at least a portion of the off stateis to be electrically biased so as to improve the linearity of the atleast one field effect transistor relative to the at least one fieldeffect transistor not electrically biased to have a voltage levelsubstantially more negative than the lowest voltage level of thefollowing: ground, the DC voltage level of the source of the at leastone field effect transistor, and the DC voltage level of the drain ofthe at least one field effect transistor.
 17. The communication deviceof claim 11, wherein the at least one field effect transistor during theat least a portion of the off state is to be electrically biased so asto reduce non-linear harmonic and/or intermodulation distortion of RFsignals to be propagated by the communication device, reduction via theat least one field effect transistor being relative to the at least onefield effect transistor not electrically biased to have a voltage levelsubstantially more negative than the lowest voltage level of thefollowing: ground, the DC voltage level of the source of the at leastone field effect transistor, and the DC voltage level of the drain ofthe at least one field effect transistor.
 18. The communication deviceof claim 17, wherein the power of a third harmonic of the RF signals tobe propagated via the RF switch is to be lower than −30 dBm at anoperating power of +35 dBm power.
 19. The communication device of claim11, wherein the at least one field effect transistor is included in astack of field effect transistors included in an RF switch, the RFswitch being included in the at least one integrated circuit chip. 20.The communication device of claim 11, wherein, during the at least aportion of the off state, the body of the at least one field effecttransistor is to be electrically coupled to at least two accumulatedcharge sinks.
 21. An RF front-end comprising: at least one integratedcircuit chip and other RF front-end components; the at least oneintegrated circuit chip and other RF front-end components included inthe RF front-end, the at least one integrated circuit chip furtherincluding at least one field effect transistor, the at least one fieldeffect transistor includes a gate, a drain, a source, and a body;wherein, during at least a portion of the off state, the body of the atleast one field effect transistor is to be electrically biased to have avoltage level substantially more negative than the lowest voltage levelof the following: ground, the DC voltage level of the source of the atleast one field effect transistor, and the DC voltage level of the drainof the at least one field effect transistor.
 22. The RF module of claim21, wherein the at least one field effect transistor comprises a N-typemetal oxide semiconductor (NMOS) field effect transistor.
 23. The RFfront-end of claim 21, wherein the at least one field effect transistoris implemented in a silicon on insulator technology.
 24. The RFfront-end of claim 21, wherein the voltage level substantially morenegative than the lowest voltage level is more than one volt morenegative than the lowest voltage level.
 25. The RF front-end of claim21, wherein the body of the at least one field effect transistor iselectrically coupled to an accumulated charge sink.
 26. The RF front-endof claim 21, wherein the at least one field effect transistor during theat least a portion of the off state is to be electrically biased so asto improve the linearity of the at least one field effect transistorrelative to the at least one field effect transistor not electricallybiased to have a voltage level substantially more negative than thelowest voltage level of the following: ground, the DC voltage level ofthe source of the at least one field effect transistor, and the DCvoltage level of the drain of the at least one field effect transistor.27. The RF front-end of claim 21, wherein the at least one field effecttransistor during the at least a portion of the off state is to beelectrically biased so as to reduce non-linear harmonic and/orintermodulation distortion of RF signals to be propagated by the RFfront-end, reduction via the at least one field effect transistorrelative to the at least one field effect transistor not electricallybiased to have a voltage level substantially more negative than thelowest voltage level of the following: ground, the DC voltage level ofthe source of the at least one field effect transistor, and the DCvoltage level of the drain of the at least one field effect transistor.28. The RF front-end of claim 27, wherein the power of a third harmonicof the RF signals to be propagated via the RF front-end is to be lowerthan −30 dBm at an operating power of +35 dBm power.
 29. The RFfront-end of claim 21, wherein the at least one field effect transistoris included in a stack of field effect transistors included in the RFswitch, the RF switch being incorporated in the at least one integratedcircuit chip.
 30. The RF front-end of claim 21, wherein the body of theat least one field effect transistor is electrically coupled to at leasttwo accumulated.